MECANISMO DE ADAPTACIÓN PARA EL SENSOR DE PESO
MÓDULO DE IMPRESIÓN 3D PARA LA PANTALLA GLCD Y LA PCB
CARGOS POR EXCESO DE EQUIPAJE-LAN PERÚ
PATENTES DE MALETAS
DATOS DE LA CELDA DE CARGA
Single and Dual-Supply, Rail-to-Rail,
Low Cost Instrumentation Amplifier
Data Sheet AD623
FEATURES GENERAL DESCRIPTION
Easy to use The AD623 is an integrated, single- or dual-supply instrumentation
Rail-to-rail output swing amplifier that delivers rail-to-rail output swing using supply
Input voltage range extends 150 mV below ground voltages from 3 V to 12 V. The AD623 offers superior user
(single supply) flexibility by allowing single gain set resistor programming and by
Low power, 550 μA maximum supply current conforming to the 8-lead industry standard pinout configuration.
Gain set with one external resistor With no external resistor, the AD623 is configured for unity
Gain range: 1 to 1000 gain (G = 1), and with an external resistor, the AD623 can be
High accuracy dc performance programmed for gains of up to 1000.
0.10% gain accuracy (G = 1) The superior accuracy of the AD623 is the result of increasing
0.35% gain accuracy (G > 1) ac common-mode rejection ratio (CMRR) coincident with
Noise: 35 nV/√Hz RTI noise at 1 kHz increasing gain; line noise harmonics are rejected due to
Excellent dynamic specifications constant CMRR up to 200 Hz. The AD623 has a wide input
800 kHz bandwidth (G = 1) common-mode range and amplifies signals with common-
20 μs settling time to 0.01% (G = 10) mode voltages as low as 150 mV below ground. The AD623
APPLICATIONS maintains superior performance with dual and single polarity
Low power medical instrumentation power supplies.
Transducer interfaces Table 1. Low Power Upgrades for the AD623
Thermocouple amplifiers
Part No. Total VS (V dc) Typical IQ (μA)
Industrial process controls
AD8235 5.5 30
Difference amplifiers
AD8236 5.5 33
Low power data acquisition
AD8237 5.5 33
AD8226 36 350
AD8227 36 325
AD8420 36 85
AD8422 36 300
AD8426 36 325 (per channel)
FUNCTIONAL BLOCK DIAGRAM
+VS
2 7
–IN
50kΩ 50kΩ
– A1VDIFF
2 + –RG 50kΩ
V 6CM – R 1 A3 OUTPUTVDIFF G 8
2 50kΩ+
+RG
+IN
3 50kΩ 50kΩ 54 A2 REF
–VS
Figure 1.
Rev. E Document Feedback
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rights of third parties that may result from its use. Specifications subject to change without notice. No One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Tel: 781.329.4700 ©1997–2016 Analog Devices, Inc. All rights reserved.
Trademarks and registered trademarks are the property of their respective owners. Technical Support www.analog.com
00778-054
AD623* PRODUCT PAGE QUICK LINKS
Last Content Update: 02/23/2017
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Instrumentation Amplifier Data Sheet
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AD623 Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1 Applications Information .............................................................. 18
Applications ....................................................................................... 1 Basic Connection ....................................................................... 18
General Description ......................................................................... 1 Gain Selection ............................................................................. 18
Functional Block Diagram .............................................................. 1 Reference Terminal .................................................................... 18
Revision History ............................................................................... 2 Input and Output Offset Voltage Error ................................... 19
Specifications ..................................................................................... 3 Input Protection ......................................................................... 19
Single Supply ................................................................................. 3 RF Interference ........................................................................... 19
Grounding ................................................................................... 20 Dual Supplies ................................................................................ 5
Input Differential and Common-Mode Range vs.
Specifications Common to Dual and Single Supplies ............. 7 Supply and Gain ......................................................................... 22
Absolute Maximum Ratings ............................................................ 8 Additional Information ............................................................. 23
ESD Caution .................................................................................. 8 Evaluation Board ............................................................................ 24
Pin Configuration and Function Descriptions ............................. 9 General Description ................................................................... 24
Typical Performance Characteristics ........................................... 10 Outline Dimensions ....................................................................... 25
Theory of Operation ...................................................................... 17 Ordering Guide .......................................................................... 26
REVISION HISTORY
6/2016—Rev. D to Rev. E 7/2008—Rev. C to Rev. D
Changes to Features Section, General Description Section, Updated Format .................................................................. Universal
and Figure 1 ....................................................................................... 1 Changes to Features Section and General Description Section .. 1
Deleted Connection Diagram Section ........................................... 1 Changes to Table 3 ............................................................................. 6
Added Functional Block Diagram Section and Table 1; Changes to Figure 40 ...................................................................... 14
Renumbered Sequentially ................................................................ 1 Changes to Theory of Operation Section.................................... 15
Changes to Single Supply Section ................................................... 3 Changes to Figure 42 and Figure 43............................................. 16
Changes to Table 3 ............................................................................ 6 Changes to Table 7 .......................................................................... 19
Changed Both Dual and Single Supplies Section to Updated Outline Dimensions ....................................................... 22
Specifications Common to Dual and Single Supplies Section ... 7 Changes to Ordering Guide .......................................................... 23
Changes to Table 5 ............................................................................ 8
Added Pin Configuration and Function Descriptions Section, 9/1999—Rev. B to Rev. C
Figure 2, and Table 6; Renumbered Sequentially ......................... 9
Changes to Figure 5 Caption, Figure 6 Caption, and
Figure 8 Caption ............................................................................. 10
Changes to Figure 17 Caption through Figure 20 Caption ....... 11
Changes to Figure 21 Caption through Figure 26 Caption ....... 12
Changes to Figure 27 Caption and Figure 28 Caption .............. 13
Changes to Theory of Operation Section .................................... 17
Changes to Basic Connection Section ......................................... 18
Changes to Input and Output Offset Voltage Error Section, and
Input Protection Section ................................................................ 19
Added Additional Information Section ....................................... 23
Added Evaluation Board Section and Figure 56 ........................ 24
Updated Outline Dimensions ....................................................... 25
Changes to Ordering Guide .......................................................... 26
Rev. E | Page 2 of 26
Data Sheet AD623
SPECIFICATIONS
SINGLE SUPPLY
Typical at 25°C, single supply, +VS = 5 V, −VS = 0 V, and RL = 10 kΩ, unless otherwise noted.
Table 2.
Test Conditions/ AD623A AD623ARM AD623B
Parameter Comments Min Typ Max Min Typ Max Min Typ Max Unit
GAIN G = 1 + (100 k/RG)
Gain Range 1 1000 1 1000 1 1000
Gain Error1 G1 VOUT =
0.05 V to 3.5 V
G > 1 VOUT =
0.05 V to 4.5 V
G = 1 0.03 0.10 0.03 0.10 0.03 0.05 %
G = 10 0.10 0.35 0.10 0.35 0.10 0.35 %
G = 100 0.10 0.35 0.10 0.35 0.10 0.35 %
G = 1000 0.10 0.35 0.10 0.35 0.10 0.35 %
Nonlinearity G1 VOUT =
0.05 V to 3.5 V
G > 1 VOUT =
0.05 V to 4.5 V
G = 1 to 1000 50 50 50 ppm
Gain vs. Temperature
G = 1 5 10 5 10 5 10 ppm/°C
G > 11 50 50 50 ppm/°C
VOLTAGE OFFSET Total RTI error =
VOSI + VOSO/G
Input Offset, VOSI 25 200 200 500 25 100 μV
Over Temperature 350 650 160 μV
Average Temperature 0.1 2 0.1 2 0.1 1 μV/°C
Coefficient (Tempco)
Output Offset, VOSO 200 1000 500 2000 200 500 μV
Over Temperature 1500 2600 1100 μV
Average Tempco 2.5 10 2.5 10 2.5 10 μV/°C
Offset Referred to the
Input vs. Supply (PSR)
G = 1 80 100 80 100 80 100 dB
G = 10 100 120 100 120 100 120 dB
G = 100 120 140 120 140 120 140 dB
G = 1000 120 140 120 140 120 140 dB
INPUT CURRENT
Input Bias Current 17 25 17 25 17 25 nA
Over Temperature 27.5 27.5 27.5 nA
Average Tempco 25 25 25 pA/°C
Input Offset Current 0.25 2 0.25 2 0.25 2 nA
Over Temperature 2.5 2.5 2.5 nA
Average Tempco 5 5 5 pA/°C
INPUT
Input Impedance
Differential 2||2 2||2 2||2 GΩ||pF
Common-Mode 2||2 2||2 2||2 GΩ||pF
Input Voltage Range2 VS = 3 V to 12 V (−VS) − (+VS) − (−VS) − (+VS) − (−VS) − (+VS) − V
0.15 1.5 0.15 1.5 0.15 1.5
Rev. E | Page 3 of 26
AD623 Data Sheet
Test Conditions/ AD623A AD623ARM AD623B
Parameter Comments Min Typ Max Min Typ Max Min Typ Max Unit
Common-Mode Rejection
at 60 Hz with 1 kΩ
Source Imbalance
G = 1 VCM = 0 V to 3 V 70 80 70 80 77 86 dB
G = 10 VCM = 0 V to 3 V 90 100 90 100 94 100 dB
G = 100 VCM = 0 V to 3 V 105 110 105 110 105 110 dB
G = 1000 VCM = 0 V to 3 V 105 110 105 110 105 110 dB
OUTPUT
Output Swing RL = 10 kΩ 0.01 (+VS) − 0.01 (+VS) − 0.01 (+VS) − V
0.5 0.5 0.5
RL = 100 kΩ 0.01 (+VS) − 0.01 (+VS) − 0.01 (+VS) − V
0.15 0.15 0.15
DYNAMIC RESPONSE
Small Signal −3 dB BW
G = 1 800 800 800 kHz
G = 10 100 100 100 kHz
G = 100 10 10 10 kHz
G = 1000 2 2 2 kHz
Slew Rate 0.3 0.3 0.3 V/μs
Settling Time to 0.01% VS = 5 V
G = 1 Step size: 3.5 V 30 30 30 μs
G = 10 Step size: 4 V, 20 20 20 μs
VCM = 1.8 V
1 Does not include effects of external resistor, RG.
2 One input grounded. G = 1.
Rev. E | Page 4 of 26
Data Sheet AD623
DUAL SUPPLIES
Typical at 25°C dual supply, VS = ±5 V, and RL = 10 kΩ, unless otherwise noted.
Table 3.
Test Conditions/ AD623A AD623ARM AD623B
Parameter Comments Min Typ Max Min Typ Max Min Typ Max Unit
GAIN G = 1 + (100 k/RG)
Gain Range 1 1000 1 1000 1 1000
Gain Error1 G1 VOUT =
−4.8 V to +3.5 V
G > 1 VOUT =
0.05 V to 4.5 V
G = 1 0.03 0.10 0.03 0.10 0.03 0.05 %
G = 10 0.10 0.35 0.10 0.35 0.10 0.35 %
G = 100 0.10 0.35 0.10 0.35 0.10 0.35 %
G = 1000 0.10 0.35 0.10 0.35 0.10 0.35 %
Nonlinearity G1 VOUT =
−4.8 V to +3.5 V
G > 1 VOUT =
−4.8 V to +4.5 V
G = 1 to 1000 50 50 50 ppm
Gain vs. Temperature
G = 1 5 10 5 10 5 10 ppm/°C
G > 11 50 50 50 ppm/°C
VOLTAGE OFFSET Total RTI error =
VOSI + VOSO/G
Input Offset, VOSI 25 200 200 500 25 100 μV
Over Temperature 350 650 160 μV
Average Tempco 0.1 2 0.1 2 0.1 1 μV/°C
Output Offset, VOSO 200 1000 500 2000 200 500 μV
Over Temperature 1500 2600 1100 μV
Average Tempco 2.5 10 2.5 10 2.5 10 μV/°C
Offset Referred to the
Input vs. Supply (PSR)
G = 1 80 100 80 100 80 100 dB
G = 10 100 120 100 120 100 120 dB
G = 100 120 140 120 140 120 140 dB
G = 1000 120 140 120 140 120 140 dB
INPUT CURRENT
Input Bias Current 17 25 17 25 17 25 nA
Over Temperature 27.5 27.5 27.5 nA
Average Tempco 25 25 25 pA/°C
Input Offset Current 0.25 2 0.25 2 0.25 2 nA
Over Temperature 2.5 2.5 2.5 nA
Average Tempco 5 5 5 pA/°C
INPUT
Input Impedance
Differential 2||2 2||2 2||2 GΩ||pF
Common-Mode 2||2 2||2 2||2 GΩ||pF
Input Voltage Range2 VS = (−VS) – (+VS) – (−VS) – (+VS) – (−VS) – (+VS) – V
+2.5 V to ±6 V 0.15 1.5 0.15 1.5 0.15 1.5
Rev. E | Page 5 of 26
AD623 Data Sheet
Test Conditions/ AD623A AD623ARM AD623B
Parameter Comments Min Typ Max Min Typ Max Min Typ Max Unit
Common-Mode Rejection
at 60 Hz with 1 kΩ
Source Imbalance
G = 1 VCM = 70 80 70 80 77 86 dB
+3.5 V to −5.15 V
G = 10 VCM = 90 100 90 100 94 100 dB
+3.5 V to −5.15 V
G = 100 VCM = 105 110 105 110 105 110 dB
+3.5 V to −5.15 V
G = 1000 VCM = 105 110 105 110 105 110 dB
+3.5 V to −5.15 V
OUTPUT
Output Swing RL = 10 kΩ, (−VS) + (+VS) − (−VS) + (+VS) − (−VS) + (+VS) − V
VS = ±5 V 0.2 0.5 0.2 0.5 0.2 0.5
RL = 100 kΩ (−VS) + (+VS) − (−VS) + (+VS) − (−VS) + (+VS) − V
0.05 0.15 0.05 0.15 0.05 0.15
DYNAMIC RESPONSE
Small Signal −3 dB
Bandwidth
G = 1 800 800 800 kHz
G = 10 100 100 100 kHz
G = 100 10 10 10 kHz
G = 1000 2 2 2 kHz
Slew Rate 0.3 0.3 0.3 V/μs
Settling Time to 0.01% VS = ±5 V, 5 V step
G = 1 30 30 30 μs
G = 10 20 20 20 μs
1 Does not include effects of external resistor, RG.
2 One input grounded. G = 1.
Rev. E | Page 6 of 26
Data Sheet AD623
SPECIFICATIONS COMMON TO DUAL AND SINGLE SUPPLIES
Table 4.
Test Conditions/ AD623A AD623ARM AD623B
Parameter Comments Min Typ Max Min Typ Max Min Typ Max Unit
NOISE
Voltage Noise, 1 kHz Total RTI noise =
e 2ni 2eno /G2
Input, Voltage Noise, eni 35 35 35 nV/√Hz
Output, Voltage Noise, eno 50 50 50 nV/√Hz
RTI, 0.1 Hz to 10 Hz
G = 1 3.0 3.0 3.0 μV p-p
G = 1000 1.5 1.5 1.5 μV p-p
Current Noise f = 1 kHz 100 100 100 fA/√Hz
0.1 Hz to 10 Hz 1.5 1.5 1.5 pA p-p
REFERENCE INPUT
RIN 100 ± 100 ± 100 ± kΩ
20% 20% 20%
IIN VIN+, VREF = 0 V 50 60 50 60 50 60 μA
Voltage Range −VS +VS −VS +VS −VS +VS V
Gain to Output 1 ± 1 ± 1 ± V
0.0002 0.0002 0.0002
POWER SUPPLY
Operating Range Dual supply ±2.5 ±6 ±2.5 ±6 ±2.5 ±6 V
Single supply 2.7 12 2.7 12 2.7 12 V
Quiescent Current Dual supply 375 550 375 550 375 550 μA
Single supply 305 480 305 480 305 480 μA
Over Temperature 625 625 625 μA
TEMPERATURE RANGE
For Specified Performance −40 +85 −40 +85 −40 +85 °C
Rev. E | Page 7 of 26
AD623 Data Sheet
ABSOLUTE MAXIMUM RATINGS
Table 5. Stresses at or above those listed under Absolute Maximum
Parameter Rating Ratings may cause permanent damage to the product. This is a
Supply Voltage 12 V stress rating only; functional operation of the product at these
Internal Power Dissipation1 650 mW or any other conditions above those indicated in the operational
Differential Input Voltage ±6 V section of this specification is not implied. Operation beyond
Output Short-Circuit Duration Indefinite the maximum operating conditions for extended periods may
Storage Temperature Range −65°C to +125°C affect product reliability.
Operating Temperature Range −40°C to +85°C
Lead Temperature (Soldering, 10 sec) 300°C ESD CAUTION
1 Specification is for device in free air:
8-Lead PDIP Package: θJA = 95°C/W
8-Lead SOIC Package: θJA = 155°C/W
8-Lead MSOP Package: θJA = 200°C/W
Rev. E | Page 8 of 26
Data Sheet AD623
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
AD623
–RG 1 8 +RG
–IN 2 7 +VS
+IN 3 6 OUTPUT
–VS 4 5 REF
TOP VIEW
(Not to Scale)
Figure 2. AD623 Pin Configuration
Table 6. Pin Function Descriptions
Pin No. Mnemonic Description
1 −RG Inverting Terminal of External Gain-Setting Resistor, RG.
2 −IN Inverting In-Amp Input.
3 +IN Noninverting In-Amp Input.
4 −VS Negative Supply Terminal.
5 REF In-Amp Output Reference Input. The voltage input establishes the common-mode voltage of the output.
6 OUTPUT In-Amp Output.
7 +VS Positive Supply Terminal.
8 +RG Noninverting Terminal of External Gain Setting Resistor, RG.
Rev. E | Page 9 of 26
00778-001
AD623 Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
At 25°C, VS = ±5 V, and RL = 10 kΩ, unless otherwise noted.
300 22
280
260 20
240 18
220 16
200
180 14
160 12
140 10
120
100 8
80 6
60 4
40
20 2
0 0
–100 –80 –60 –40 –20 0 20 40 60 80 100 120 140 –600 –500 –400 –300 –200 –100 0 100 200 300 400 500
INPUT OFFSET VOLTAGE (µV) OUTPUT OFFSET VOLTAGE (µV)
Figure 3. Typical Distribution of Input Offset Voltage, Figure 6. Typical Distribution of Output Offset Voltage,
N-8 and R-8 Package Options +VS = 5 V, −VS = 0 V, VREF = −0.125 V, N-8 and R-8 Package Options
480 210
420 180
360
150
300
120
240
90
180
120 60
60 30
0 0
–800 –600 –400 –200 0 200 400 600 800 –0.245 –0.240 –0.235 –0.230 –0.225 –0.220 –0.215 –0.210
OUTPUT OFFSET VOLTAGE (µV) INPUT OFFSET CURRENT (nA)
Figure 4. Typical Distribution of Output Offset Voltage, Figure 7. Typical Distribution for Input Offset Current,
N-8 and R-8 Package Options N-8 and R-8 Package Options
22 20
20 18
18 16
16
14
14
12
12
10
10
8 8
6 6
4 4
2 2
0 0
–80 –60 –40 –20 0 20 40 60 80 100 –0.025 –0.020 –0.015 –0.010 –0.005 0 0.005 0.010
INPUT OFFSET VOLTAGE (µV) INPUT OFFSET CURRENT (nA)
Figure 5. Typical Distribution of Input Offset Voltage, Figure 8. Typical Distribution for Input Offset Current,
+VS = 5 V, −VS = 0 V, VREF = −0.125 V, N-8 and R-8 Package Options +VS = 5 V, −VS = 0 V, VREF = −0.125 V, N-8 and R-8 Package Options
Rev. E | Page 10 of 26
UNITS UNITS UNITS
00778-005 00778-004 00778-003
UNITS UNITS UNITS
00778-008
00778-007 00778-006
Data Sheet AD623
1600 30
1400
25
1200
20
1000
800 15
600
10
400
5
200
0 0
75 80 85 90 95 100 105 110 115 120 125 130 –60 –40 –20 0 20 40 60 80 100 120 140
CMRR (dB) TEMPERATURE (°C)
Figure 9. Typical Distribution for CMRR (G = 1) Figure 12. IBIAS vs. Temperature
1k 1k
G = 1
100 100
G= 10
G= 100
G= 1000
10 10
1 10 100 1k 10k 100k 1 10 100 1k
FREQUENCY (Hz) FREQUENCY (Hz)
Figure 10. Voltage Noise Spectral Density vs. Frequency Figure 13. Current Noise Spectral Density vs. Frequency
22 20.0
21 19.5
20 19.0
19 18.5
18 18.0
17 17.5
16 17.0
15 16.5
14 16.0
–4 –2 0 2 4 –4 –3 –2 –1 0 1 2
CMV (V) CMV (V)
Figure 11. IBIAS vs. CMV Figure 14. IBIAS vs. CMV, VS = ±2.5 V
Rev. E | Page 11 of 26
I (nA) VOLTAGE NOISE SPECTRAL DENSITY (nV/ Hz RTI)BIAS UNITS
00778-011 00778-009
00778-010
IBIAS (nA) CURRENT NOISE SPECTRAL DENSITY (fA/ Hz)
IBIAS (nA)
00778-014 00778-013
00778-012
AD623 Data Sheet
120
CH1 10mV A 1s 100mV VERT
110
100
G = ×1000
90
80
G = ×100
70
60
G = ×10
50
40 G = ×1
30
1 10 100 1k 10k 100k
FREQUENCY (Hz)
Figure 15. 0.1 Hz to 10 Hz Current Noise (0.71 pA/DIV) Figure 18. CMR vs. Frequency for Various Gain Settings (G)
70
1µV/DIV 1s G = 1000
60
50
G = 100
40
30
G = 10
20
10
G = 1
0
–10
–20
–30
100 1k 10k 100k 1M
FREQUENCY (Hz)
Figure 16. 0.1 Hz to 10 Hz RTI Voltage Noise (1 DIV = 1 μV p-p) Figure 19. Gain vs. Frequency (+VS = 5 V, −VS = 0 V), VREF = 2.5 V,
for Various Gain Settings (G)
120 5
110 4
3
100 G = ×1000
2
90
G = ×100 1 VS = ±2.5V
80 0
70 –1
G = ×10
60 –2
–3
50
–4
40 G = ×1 –5
30 –6
1 10 100 1k 10k 100k –5 –4 –3 –2 –1 0 1 2 3 4 5
FREQUENCY (Hz) MAXIMUM OUTPUT VOLTAGE (V)
Figure 17. Common-Mode Rejection (CMR) vs. Frequency, +VS = 5 V, − VS = 0 Figure 20. Maximum Output Voltage vs. Common-Mode Input,
V, VREF = 2.5 V, for Various Gain Settings (G) G = 1, RL = 100 kΩ for Two Supply Voltages
Rev. E | Page 12 of 26
CMR (dB)
00778-016 00778-015
00778-017
COMMON-MODE INPUT (V) GAIN (dB) CMR (dB)
00778-020 00778-018
00778-019
Data Sheet AD623
5 140
4
120
3 G = 1000
VS = ±2.5V2 100
1 G = 100
0 80
–1 60
–2 G = 10
–3 40
G = 1
–4
20
–5
–6 0
–5 –4 –3 –2 –1 0 1 2 3 4 5 1 10 100 1k 10k 100k
MAXIMUM OUTPUT VOLTAGE (V) FREQUENCY (Hz)
Figure 21. Maximum Output Voltage vs. Common-Mode Input, Figure 24. Positive PSRR vs. Frequency
G ≥ 10, RL = 100 Ω, for Two Supply Voltages
5 140
4 120 G = 1000
100
3 G = 100
80
2
60
G = 10
1
40
G = 1
0 20
–1 0
0 1 2 3 4 5 1 10 100 1k 10k 100k
MAXIMUM OUTPUT VOLTAGE (V) FREQUENCY (Hz)
Figure 22. Maximum Output Voltage vs. Common-Mode Input, Figure 25. Positive PSRR vs. Frequency, +VS = 5V, −VS = 0 V,
G = 1, +VS = 5 V, −VS = 0 V, RL = 100 kΩ for Various Gain Settings (G)
5 140
120 G = 1000
4
G = 100
100
3
80
2 G = 10
60
G = 1
1
40
0 20
–1 0
0 1 2 3 4 5 1 10 100 1k 10k 100k
MAXIMUM OUTPUT VOLTAGE (V) FREQUENCY (Hz)
Figure 23. Maximum Output Voltage vs. Common-Mode Input, Figure 26. Negative PSRR vs. Frequency for Various Gain Settings (G)
G ≥ 10, +VS = 5 V, −VS = 0 V, RL = 100 kΩ
Rev. E | Page 13 of 26
COMMON-MODE INPUT (V) COMMON-MODE INPUT (V) COMMON-MODE INPUT (V)
00778-023 00778-022 00778-021
NEGATIVE PSRR (dB) POSITIVE PSSR (dB) POSITIVE PSSR (dB)
00778-026 00778-025 00778-024
AD623 Data Sheet
10
500µV 1V 10µs
8
6
4
VS = ±5V
VS = ±2.5V
2
0
0 20 40 60 80 100
FREQUENCY (kHz)
Figure 27. Large Signal Response, G ≤ 10 for Two Supply Voltages Figure 30. Large Signal Pulse Response and Settling Time,
G = −10 (0.250 mV = 0.01%), CL = 100 pF
1k
10mV 2V 50µs
100
10
1
1 10 100 1k
GAIN (V/V)
Figure 28. Settling Time to 0.01% vs. Gain, for a 5 V Step at Output, Figure 31. Large Signal Pulse Response and Settling Time,
CL = 100 pF G = 100, CL = 100 pF
500µV 1V 20µs
20mV 2V 500µs
Figure 29. Large Signal Pulse Response and Settling Time, Figure 32. Large Signal Pulse Response and Settling Time,
G = −1 (0.250 mV = 0.01%), CL = 100 pF G = −1000 (5 mV = 0.01%), CL = 100 pF
Rev. E | Page 14 of 26
SETTLING TIME (µs) OUTPUT VOLTAGE (V p-p)
00778-029
00778-028
00778-027
00778-032 00778-031 00778-030
Data Sheet AD623
20mV 2µs 20mV 500µs
Figure 33. Small Signal Pulse Response, G = 1, RL = 10 kΩ, CL = 100 pF Figure 36. Small Signal Pulse Response, G = 1000, RL = 10 kΩ, CL = 100 pF
20mV 5µs 200µV
1V
Figure 34. Small Signal Pulse Response, G = 10, RL = 10 kΩ, CL = 100 pF Figure 37. Gain Nonlinearity, G = −1 (50 ppm/DIV)
20mV 50µs 20µV 1V
Figure 35. Small Signal Pulse Response, G = 100, RL = 10 kΩ, CL = 100 pF Figure 38. Gain Nonlinearity, G = −10 (6 ppm/DIV)
Rev. E | Page 15 of 26
00778-035 00778-034 00778-033
00778-038 00778-037 00778-036
AD623 Data Sheet
V+
50µV 1V
(V+) –0.5
(V+) –1.5
(V+) –2.5
(V–) +0.5
V–
0 0.5 1.0 1.5 2.0
OUTPUT CURRENT (mA)
Figure 39. Gain Nonlinearity, G = −100, 15 ppm/DIV Figure 40. Output Voltage Swing vs. Output Current
Rev. E | Page 16 of 26
00778-039
OUTPUT VOLTAGE SWING (V)
00778-040
Data Sheet AD623
THEORY OF OPERATION
The AD623 is an instrumentation amplifier based on a modified The output voltage at Pin 6 is measured with respect to the
classic 3-op-amp approach, to assure single- or dual-supply potential at Pin 5. The impedance of the reference pin is 100 kΩ;
operation even at common-mode voltages at the negative supply therefore, in applications requiring voltage conversion, a small
rail. Low voltage offsets, input and output, as well as absolute resistor between Pin 5 and Pin 6 is all that is needed.
gain accuracy, and one external resistor to set the gain, make +VS
the AD623 one of the most versatile instrumentation amplifiers 7
in its class.
The input signal is applied to PNP transistors acting as voltage 2–IN
buffers and providing a common-mode signal to the input 4
–V 50kΩ 50kΩ 50kΩ
amplifiers (see Figure 41). An absolute value 50 kΩ resistor in –R 1 SG
each amplifier feedback assures gain programmability.
R 6 OTUPUT
The differential output is G
50kΩ 50kΩ 50kΩ
100 kΩ +RG 5 REF
VO 1
8 +VS
R
VC 7
G
The differential voltage is then converted to a single-ended 3
voltage using the output amplifier, which also rejects any +IN
common-mode signal at the output of the input amplifiers. 4–VS
Because the amplifiers can swing to either supply rail, as well as Figure 41. Simplified Schematic
have their common-mode range extended to below the negative Because of the voltage feedback topology of the internal op
supply rail, the range over which the AD623 can operate is further amps, the bandwidth of the in-amp decreases with increasing
enhanced (see Figure 20 and Figure 21). gain. At unity gain, the output amplifier limits the bandwidth.
Rev. E | Page 17 of 26
00778-041
AD623 Data Sheet
APPLICATIONS INFORMATION
BASIC CONNECTION The input voltage, which can be either single-ended (tie either
Figure 42 and Figure 43 show the basic connection circuits for −IN or +IN to ground) or differential, is amplified by the
the AD623. The +VS and −VS terminals are connected to the programmed gain. The output signal appears as the voltage
power supply. The supply can be either bipolar (V difference between the OUTPUT pin and the externally applied S = ±2.5 V to
±6 V) or single supply (−V voltage on the REF input. For a ground referenced output, REF S = 0 V, +VS = 3.0 V to 12 V).
Capacitively decouple power supplies close to the power pins of must be grounded.
the device. For best results, use surface-mount 0.1 μF ceramic GAIN SELECTION
chip capacitors and 10 μF electrolytic tantalum capacitors. The gain of the AD623 is programmed by the RG resistor, or
+VS more precisely, by whatever impedance appears between Pin 1
0.1µF 10µF
and Pin 8. The AD623 offers accurate gains using 0.1% to 1%
+2.5V TO +6V tolerance resistors. Table 7 shows the required values of RG for
the various gains. Note that for G = 1, the RG terminals are
RG
VIN RG OUTPUT VOUT unconnected (RG = ∞). For any arbitrary gain, RG can be
RG REF calculated by
REF (INPUT)
0.1µF 10µF RG = 100 kΩ/(G − 1)
–VS REFERENCE TERMINAL
–2.5V TO –6V The reference terminal potential defines the zero output voltage
Figure 42. Dual-Supply Basic Connection and is especially useful when the load does not share a precise
+VS
0.1µF 10µF ground with the rest of the system. It provides a direct means of
injecting a precise offset to the output. The reference terminal is
+3V TO +12V also useful when bipolar signals are being amplified because it
can be used to provide a virtual ground voltage. The voltage on
RG
V R OUTPUT V the reference terminal can be varied from −VS to +VS. IN G OUT
RG REF
REF (INPUT)
Figure 43. Single-Supply Basic Connection
Table 7. Required Values of Gain Resistors
Desired Gain 1% Standard Table Value of RG Calculated Gain Using 1% Resistors
2 100 kΩ 2
5 24.9 kΩ 5.02
10 11 kΩ 10.09
20 5.23 kΩ 20.12
33 3.09 kΩ 33.36
40 2.55 kΩ 40.21
50 2.05 kΩ 49.78
65 1.58 kΩ 64.29
100 1.02 kΩ 99.04
200 499 Ω 201.4
500 200 Ω 501
1000 100 Ω 1001
Rev. E | Page 18 of 26
00778-055 00778-042
Data Sheet AD623
INPUT AND OUTPUT OFFSET VOLTAGE ERROR RF INTERFERENCE
The offset voltage (VOS ) of the AD623 is attributed to two All instrumentation amplifiers can rectify high frequency out-
sources: those originating in the two input stages where the in- of-band signals. Once rectified, these signals appear as dc offset
amp gain is established, and those originating in the subtractor errors at the output. The circuit in Figure 45 provides good RFI
output stage. The output error is divided by the programmed suppression without reducing performance within the pass band of
gain when referred to the input. In practice, the input errors the in-amp. Resistor R1 and Capacitor C1 (and likewise, R2 and
dominate at high gain settings, whereas the output error C2) form a low-pass RC filter that has a −3 dB bandwidth equal
prevails when the gain is set at or near unity. to f = 1/(2 π R1C1). Using the component values shown, this
The V error for any given gain is calculated as follows: filter has a −3 dB bandwidth of approximately 40 kHz. The R1 OS
and R2 resistors were selected to be large enough to isolate the
Total Error Referred to Input (RTI) input of the circuit from the capacitors, but not large enough to
= Input Error + (Output Error/G) significantly increase the noise of the circuit. To preserve common-
Total Error Referred to Output (RTO) mode rejection in the pass band of the amplifier, the C1 and C2
= (Input Error × G) + Output Error capacitors must be 5% or better units, or low cost 20% units can
The RTI offset errors and noise voltages for different gains are be tested and binned to provide closely matched devices.
listed in Table 8. +VS
0.33µF 0.01µF
INPUT PROTECTION R1 C1
4.02kΩ 1000pF
Internal supply-referenced clamping diodes allow the input, 1% 5%–IN
reference, output, and gain terminals of the AD623 to safely R2 C3 R AD623 V
withstand overvoltages of 0.3 V above or below the supplies. 4.02kΩ 0.047µF G OUT1% REFERENCE
This overvoltage protection is true at all gain settings and when +IN C2
cycling power on and off. Overvoltage protection is particularly 1000pF5% 0.33µF 0.01µF
important because the signal source and amplifier may be
powered separately. +VNOTES: S
1. LOCATE C1 TO C3 AS CLOSE TO THE INPUT PINS AS POSSIBLE.
If the overvoltage is expected to exceed this value, the current Figure 45. Circuit to Attenuate RF Interference
through these diodes must be limited to about 10 mA using
external current limiting resistors (see Figure 44). The size of Capacitor C3 is needed to maintain common-mode rejection at
this resistor is defined by the supply voltage and the required low frequencies. R1/R2 and C1/C2 form a bridge circuit whose
overvoltage protection. output appears across the input pins of the in-amp. Any mismatch
between C1 and C2 unbalances the bridge and reduces the
+VS common-mode rejection. C3 ensures that any RF signals are
I = 10mA MAX common mode (the same on both in-amp inputs) and are not
VOVER AD623
RLIM applied differentially. This second low-pass network, R1 + R2 and
RG OUTPUT C3, has a −3 dB frequency equal to 1/(2π(R1 + R2)(C3)). Using
VOVER RLIM VOVER –VS + 0.7V a C3 value of 0.047 μF, the −3 dB signal bandwidth of this circuit RLIM = 10mA is approximately 400 Hz. The typical dc offset shift over frequency
–VS is less than 1.5 μV, and the RF signal rejection of the circuit is
Figure 44. Input Protection better than 71 dB. The 3 dB signal bandwidth of this circuit can
be increased to 900 Hz by reducing R1 and R2 to 2.2 kΩ. The
performance is similar to using 4 kΩ resistors, except that the
circuitry preceding the in-amp must drive a lower impedance load.
Table 8. RTI Error Sources
Maximum Total Input Offset Error (μV) Maximum Total Input Offset Drift (μV/°C) Total Input Referred Noise (nV/√Hz)
Gain AD623A AD623B AD623A AD623B AD623A AD623B
1 1200 600 12 11 62 62
2 700 350 7 6 45 45
5 400 200 4 3 38 38
10 300 150 3 2 35 35
20 250 125 2.5 1.5 35 35
50 220 110 2.2 1.2 35 35
100 210 105 2.1 1.1 35 35
1000 200 100 2 1 35 35
Rev. E | Page 19 of 26
00778-043
00778-044
AD623 Data Sheet
The circuit in Figure 45 must be built using a printed circuit GROUNDING
board (PCB) with a ground plane on both sides. All component Because the AD623 output voltage is developed with respect
leads must be as short as possible. The R1 and R2 resistors can to the potential on the reference terminal, many grounding
be common 1% metal film units; however, the C1 and C2 problems can be solved by simply tying the REF pin to the
capacitors must be ±5% tolerance devices to avoid degrading appropriate local ground. The REF pin must, however, be tied
the common-mode rejection of the circuit. Either the to a low impedance point for optimal CMR.
traditional 5% silver mica units or Panasonic ±2% PPS film
capacitors are recommended. The use of ground planes is recommended to minimize the
impedance of ground returns (and hence the size of dc errors).
In many applications, shielded cables are used to minimize noise; To isolate low level analog signals from a noisy digital environment,
for best CMR over frequency, the shield must be properly driven. many data acquisition components have separate analog and digital
Figure 46 shows an active guard driver that is configured to ground returns (see Figure 47). All ground pins from mixed signal
improve ac common-mode rejection by bootstrapping the components, such as analog-to-digital converters (ADCs), must
capacitances of input cable shields, thus minimizing the be returned through the high quality analog ground plane.
capacitance mismatch between the inputs. Maximum isolation between analog and digital is achieved by
+VS connecting the ground planes back at the supplies. The digital
–IN return currents from the ADC that flow in the analog ground
2
R 7 plane, in general, have a negligible effect on noise performance. G 1
100Ω 2
AD8031 AD623 6 OUTPUT If there is only a single power supply available, it must be shared
RG
8 5 by both digital and analog circuitry. Figure 48 shows how to 2
3 4 REF minimize interference between the digital and analog circuitry.
+IN As in the previous case, use separate analog and digital ground
–VS planes (reasonably thick traces can be used as an alternative to a
Figure 46. Common-Mode Shield Driver digital ground plane). These ground planes must be connected at
the ground pin of the power supply. Run separate traces from the
power supply to the supply pins of the digital and analog circuits.
Ideally, each device has its own power supply trace, but these can
be shared by a number of devices, as long as a single trace is not
used to route current to both digital and analog circuitry.
ANALOG POWER SUPPLY DIGITAL POWER SUPPLY
+5V –5V GND GND +5V
0.1µF 0.1µF 0.1µF 0.1µF
7
2 1 6 14
4 VDD AGND DGND 12 AGND VDD
AD623 6 4 VIN1 ADC MICROPROCESSOR
3 5 3 V AD7892-2IN2
Figure 47. Optimal Grounding Practice for a Bipolar Supply Environment with Separate Analog and Digital Supplies
POWER SUPPLY
+5V GND
0.1µF
0.1µF
0.1µF
7
2 1 6 14
4 VDD AGND DGND 12 AGND VDD
AD623 6 4 VIN1 ADC MICROPROCESSOR
3 5 AD7892-2
Figure 48. Optimal Ground Practice in a Single-Supply Environment
Rev. E | Page 20 of 26
00778-045
00778-047
00778-046
Data Sheet AD623
Ground Returns for Input Bias Currents Output Buffering
Input bias currents are those dc currents that must flow to bias The AD623 is designed to drive loads of 10 kΩ or greater. If the
the input transistors of an amplifier. These are usually transistor load is less than this value, the output of the AD623 must be
base currents. When amplifying floating input sources, such as buffered with a precision single-supply op amp, such as the
transformers or ac-coupled sources, there must be a direct dc OP113. This op amp can swing from 0 V to 4 V on its output
path into each input so that the bias current can flow. Figure 49, while driving a load as small as 600 Ω. Table 9 summarizes the
Figure 50, and Figure 51 show how a bias current path can be performance of some buffer op amps.
provided for the cases of transformer coupling, thermocouple, 5V
and capacitive ac coupling. In dc-coupled resistive bridge 0.1µF 5V
applications, providing this path is generally not necessary 0.1µF
because the bias current simply flows from the bridge supply
through the bridge into the amplifier. However, if the impedances VIN RG AD623
that the two inputs see are large and differ by a large amount OP113 VOUTREFERENCE
(>10 kΩ), the offset current of the input stage causes dc errors
proportional with the input offset voltage of the amplifier.
Figure 52. Output Buffering
+VS
–IN
2 Table 9. Buffering Options
1 7 Op Amp Description
RG AD623 6 OUTPUT OP113 Single-supply, high output current
8 5
REF OP191 Rail-to-rail input and output, low supply current 3 4
+IN
LOAD
–VS Single-Supply Data Acquisition System TO POWER
SUPPLY
GROUND Interfacing bipolar signals to single-supply ADCs presents a
Figure 49. Ground Returns for Bias Currents with Transformer-Coupled Inputs challenge. The bipolar signal must be mapped into the input
range of the ADC. Figure 53 shows how this translation can be
+VS achieved.
–IN
2 5V
1 7 5V 5V 0.1µF
RG AD623 6 OTUPUT 0.1µF
8 5
REF
3 4 R AD7776+IN ±10mV GLOAD 1.02kΩ AD623 AIN
–VS TO POWER REFERENCE
SUPPLY
GROUND REFOUT
Figure 50. Ground Returns for Bias Currents with Thermocouple Inputs REFIN
+VS
–IN
2 Figure 53. A Single-Supply Data Acquisition System
1 7
The bridge circuit is excited by a 5 V supply. The full-scale output
RG AD623 6 OUTPUT voltage from the bridge (±10 mV) therefore has a common-mode
8 5
REF level of 2.5 V. The AD623 removes the common-mode component
3 4
+IN and amplifies the input signal by a factor of 100 (RGAIN = 1.02 kΩ),
LOAD
100kΩ 100kΩ –VS TO POWER which results in an output signal of ±1 V. To prevent this signal
SUPPLY
GROUND from running into the ground rail of the AD623, the voltage on
Figure 51. Ground Returns for Bias Currents with AC-Coupled Inputs the REF pin must be raised to at least 1 V. In this example, the 2 V
reference voltage from the AD7776 ADC biases the output voltage
of the AD623 to 2 V ± 1 V, which corresponds to the input range
of the ADC.
Rev. E | Page 21 of 26
00778-050 00778-049 00778-048
00778-051
00778-052
AD623 Data Sheet
Amplifying Signals with Low Common-Mode Voltage equations, the maximum and minimum input common-mode
Because the common-mode input range of the AD623 extends voltages are given by the following equations:
0.1 V below ground, it is possible to measure small differential VCMMAX = V+ − 0.7 V − VDIFF × Gain/2
signals which have low or no common-mode component. VCMMIN = V− − 0.590 V + VDIFF × Gain/2
Figure 54 shows a thermocouple application where one side of
the J-type thermocouple is grounded. These equations can be rearranged to give the maximum possible
5V differential voltage (positive or negative) for a particular common-
mode voltage, gain, and power supply. Because the signals on
0.1µF
A1 and A2 can clip on either rail, the maximum differential
voltage is the lesser of the two equations.
J-TYPE RG
THERMOCOUPLE 1.02kΩ AD623 OUTPUT |VDIFFMAX| = 2 (V+ − 0.7 V − VCM)/Gain
REF |V
2V DIFFMAX
| = 2 (VCM − V− +0.590 V)/Gain
However, the range on the differential input voltage range is
Figure 54. Amplifying Bipolar Signals with Low Common-Mode Voltage also constrained by the output swing. Therefore, the range of
Over a temperature range of −200°C to +200°C, the J-type thermo- VDIFF may need to be lower according the following equation:
couple delivers a voltage ranging from −7.890 mV to +10.777 mV. Input Range ≤ Available Output Swing/Gain
A programmed gain on the AD623 of 100 (RG = 1.02 kΩ) and a For a bipolar input voltage with a common-mode voltage that is
voltage on the REF pin of 2 V result in the output voltage ranging roughly half way between the rails, VDIFFMAX is half the value that
from 1.110 V to 3.077 V relative to ground. the previous equations yield because the REF pin is at midsupply.
INPUT DIFFERENTIAL AND COMMON-MODE Note that the available output swing is given for different supply
RANGE vs. SUPPLY AND GAIN conditions in the Specifications section.
Figure 55 shows a simplified block diagram of the AD623. The The equations can be rearranged to give the maximum gain for
voltages at the outputs of Amplifier A1 and Amplifier A2 are a fixed set of input conditions. The maximum gain is the lesser
given by of the two equations.
VA2 = VCM + VDIFF/2 + 0.6 V + VDIFF × RF/RG GainMAX = 2 (V+ − 0.7 V − VCM)/VDIFF
= VCM + 0.6 V + VDIFF × Gain/2 GainMAX = 2 (VCM − V− +0.590 V)/VDIFF
VA1 = VCM − VDIFF/2 + 0.6 V + VDIFF × RF/RG Again, it is recommended that the resulting gain times the input
= VCM + 0.6 V − VDIFF × Gain/2 range is less than the available output swing. If this is not the case,
+VS the maximum gain is given by
7
GainMAX = Available Output Swing/Input Range
2 A1 Also for bipolar inputs (that is, input range = 2 VDIFF), the –IN maximum gain is half the value yielded by the previous equations
4 RF
– 1–V 50kΩ 50kΩ 50kΩV because the REF pin must be at midsupply. DIFF S
2 +
6 The maximum gain and resulting output swing for different input
GAIN
R A3 OUTPUT conditions is given in Table 10. Output voltages are referenced to
V GCM RF
8 50kΩ 50kΩ 50kΩ 5 the voltage on the REF pin.
+V REFS For the purposes of computation, it is necessary to break down the
V –DIFF 7
2 input voltage into its differential and common-mode components. + A2
Therefore, when one of the inputs is grounded or at a fixed
+IN 3 voltage, the common-mode voltage changes as the differential
4 voltage changes. Take the case of the thermocouple amplifier in
–VS Figure 54. The inverting input on the AD623 is grounded;
Figure 55. Simplified Block Diagram therefore, when the input voltage is −10 mV, the voltage on the
The voltages on these internal nodes are critical in determining noninverting input is −10 mV. For the purpose of the signal
whether the output voltage is clipped. The VA1 and VA2 voltages swing calculations, this input voltage must be composed of a
can swing from approximately 10 mV above the negative supply common-mode voltage of −5 mV (that is, (+IN + −IN)/2) and
(V− or ground) to within approximately 100 mV of the positive a differential input voltage of −10 mV (that is, +IN − −IN).
rail before clipping occurs. Based on this and from the previous
Rev. E | Page 22 of 26
00778-053
00778-055
Data Sheet AD623
Table 10. Maximum Attainable Gain and Resulting Output Swing for Different Input Conditions
Closest 1%
VCM VDIFF REF Pin Supply Voltages Maximum Gain Gain Resistor Resulting Gain Output Swing
0 V ±10 mV 2.5 V +5 V 118 866 Ω 116 ±1.2 V
0 V ±100 mV 2.5 V +5 V 11.8 9.31 kΩ 11.7 ±1.1 V
0 V ±10 mV 0 V ±5 V 490 205 Ω 488 ±4.8 V
0 V ±100 mV 0 V ±5 V 49 2.1 kΩ 48.61 ±4.8 V
0 V ±1 V 0 V ±5 V 4.9 26.1 kΩ 4.83 ±4.8 V
2.5 V ±10 mV 2.5 V +5 V 242 422 Ω 238 ±2.3 V
2.5 V ±100 mV 2.5 V +5 V 24.2 4.32 kΩ 24.1 ±2.4 V
2.5 V ±1 V 2.5 V +5 V 2.42 71.5 kΩ 2.4 ±2.4 V
1.5 V ±10 mV 1.5 V +3 V 142 715 Ω 141 ±1.4 V
1.5 V ±100 mV 1.5 V +3 V 14.2 7.68 kΩ 14 ±1.4 V
0 V ±10 mV 1.5 V +3 V 118 866 Ω 116 ±1.1 V
0 V ±100 mV 1.5 V +3 V 11.8 9.31 kΩ 11.74 ±1.1 V
ADDITIONAL INFORMATION For additional information on in-amps, refer to the following:
For an updated design of the AD623, see the AD8223. MT-061. Instrumentation Amplifier (In-Amp) Basics.
For a selection guide to all Analog Devices instrumentation Analog Devices, Inc.
amplifiers, see the Instrumentation Amplifiers page on the MT-070. In-Amp Input RFI Protection. Analog Devices, Inc.
Analog Devices website at www.analog.com. Counts, Lew and Charles Kitchen. A Designer's Guide to Instru-
mentation Amplifiers. 3rd edition. Analog Devices, Inc., 2006.
Rev. E | Page 23 of 26
AD623 Data Sheet
EVALUATION BOARD
GENERAL DESCRIPTION
The EVAL-INAMP-62RZ can be used to evaluate the AD620,
AD621, AD622, AD623, AD627, AD8223, and AD8225
instrumentation amplifiers. In addition to the basic in-amp
connection, circuit options enable the user to adjust the offset
voltage, apply an output reference, or provide shield drivers
with user supplied components. The board is shipped with an
assortment of instrumentation amplifier ICs in the legacy SOIC
pinout, such as the AD620, AD621, AD622, AD623, AD8223,
and AD8225. The board also has an alternative footprint for a
through-hole, 8-lead PDIP.
Figure 56 shows a photograph of the evaluation boards for all Figure 56. Evaluation Boards for Analog Devices In-Amps
Analog Devices instrumentation amplifiers. For additional
information, see the EVAL-INAMP user guide (UG-261).
Rev. E | Page 24 of 26
00778-056
Data Sheet AD623
OUTLINE DIMENSIONS
0.400 (10.16)
0.365 (9.27)
0.355 (9.02)
8 5 0.280 (7.11)
0.250 (6.35)
1 4 0.240 (6.10) 0.325 (8.26)
0.310 (7.87)
0.100 (2.54) 0.300 (7.62)
BSC 0.060 (1.52) 0.195 (4.95)
0.210 (5.33) MAX 0.130 (3.30)
MAX
0.015 0.115 (2.92)
0.150 (3.81) (0.38) 0.015 (0.38)
0.130 (3.30) MIN GAUGE
0.115 (2.92) PLANE 0.014 (0.36)SEATING
PLANE 0.010 (0.25)
0.022 (0.56) 0.008 (0.20)
0.018 (0.46) 0.005 (0.13)
0.430 (10.92)
MIN MAX
0.014 (0.36)
0.070 (1.78)
0.060 (1.52)
0.045 (1.14)
COMPLIANT TO JEDEC STANDARDS MS-001
CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.
Figure 57. 8-Lead Plastic Dual In-Line Package [PDIP]
Narrow Body (N-8)
Dimensions shown in inches and (millimeters)
5.00 (0.1968)
4.80 (0.1890)
8 5
4.00 (0.1574) 6.20 (0.2441)
3.80 (0.1497) 1 4 5.80 (0.2284)
1.27 (0.0500) 0.50 (0.0196)
BSC 1.75 (0.0688) 45°0.25 (0.0099)
0.25 (0.0098) 1.35 (0.0532) 8°
0.10 (0.0040) 0°
COPLANARITY 0.51 (0.0201)
0.10
SEATING 0.31 (0.0122)
1.27 (0.0500)
0.25 (0.0098) 0.40 (0.0157)
PLANE 0.17 (0.0067)
COMPLIANT TO JEDEC STANDARDS MS-012-AA
CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 58. 8-Lead Standard Small Outline Package [SOIC_N]
Narrow Body (R-8)
Dimensions shown in millimeters and (inches)
Rev. E | Page 25 of 26
012407-A
070606-A
AD623 Data Sheet
3.20
3.00
2.80
8 5 5.15
3.20 4.90
3.00 4.65
2.80 1 4
PIN 1
IDENTIFIER
0.65 BSC
0.95 15° MAX
0.85 1.10 MAX
0.75
0.15 0.80
0.05 0.40
6° 0.23 0.55
0° 0.09 0.40
COPLANARITY 0.25
0.10
COMPLIANT TO JEDEC STANDARDS MO-187-AA
Figure 59. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
ORDERING GUIDE
Temperature Package
Model1 Range Package Description Option Branding
AD623ANZ −40°C to +85°C 8-Lead Plastic Dual In-Line Package [PDIP] N-8
AD623AR −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
AD623AR-REEL7 −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N], 7" Tape and Reel R-8
AD623ARZ −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
AD623ARZ-R7 −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N], 7" Tape and Reel R-8
AD623ARZ-RL −40°C to +85°C 8-Lead SOIC, 13" Tape and Reel R-8
AD623ARMZ −40°C to +85°C 8-Lead Mini Small Outline Package [MSOP] RM-8 J0A
AD623ARMZ-REEL −40°C to +85°C 8-Lead Mini Small Outline Package [MSOP], 13" Tape and Reel RM-8 J0A
AD623ARMZ-REEL7 −40°C to +85°C 8-Lead Mini Small Outline Package [MSOP], 7" Tape and Reel RM-8 J0A
AD623BNZ −40°C to +85°C 8-Lead Plastic Dual In-Line Package [PDIP] N-8
AD623BRZ −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N] R-8
AD623BRZ-R7 −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N], 7" Tape and Reel R-8
AD623BRZ-RL −40°C to +85°C 8-Lead Standard Small Outline Package [SOIC_N], 13" Tape and Reel R-8
EVAL-INAMP-62RZ Evaluation Board
1 Z = RoHS Compliant Part.
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Arduino Nano (V2.3)
User Manual
Released under the Creative Commons Attribution Share-Alike 2.5 License
http://creativecommons.org/licenses/by-sa/2.5/
More information:
www.arduino.cc Rev. 2.3
Arduino Nano Pin Layout
!
D1/TX (1) (30) VIN
D0/RX (2) (29) GND
RESET (3) (28) RESET
GND (4) (27) +5V
D2 (5) (26) A0
D3 (6) (25) A1
D4 (7) (24) A2
D5 (8) (23) A3
D6 (9) (22) A4
D7 (10) (21) A5
D8 (11) (20) A6
D9 (12) (19) A7
D10 (13) (18) AREF
D11 (14) (17) 3V3
D12 (15) (16) D13
Pin No. Name Type Description
1-2, 5-16 D0-D13 I/O Digital input/output port 0 to 13
3, 28 RESET Input Reset (active low)
4, 29 GND PWR Supply ground
17 3V3 Output +3.3V output (from FTDI)
18 AREF Input ADC reference
19-26 A7-A0 Input Analog input channel 0 to 7
27 +5V Output or +5V output (from on-board regulator) or
Input +5V (input from external power supply)
30 VIN PWR Supply voltage
!
!
!
!
!
!
Arduino Nano Mechanical Drawing
!
Arduino Nano Bill of Material
Item!Number! Qty.! Ref.!Dest.! Description! Mfg.!P/N! MFG! Vendor!P/N! Vendor!
Capacitor,!0.1uF!50V!10%!
1! 5! C1,C3,C4,C7,C9! Ceramic!X7R!0805! C0805C104K5RACTU! Kemet! 80"C0805C104K5R! Mouser!
Capacitor,!4.7uF!10V!10%!
2! 3! C2,C8,C10! Tantalum!Case!A! T491A475K010AT! Kemet! 80"T491A475K010! Mouser!
Capacitor,!18pF!50V!5%!
3! 2! C5,C6! Ceramic!NOP/COG!0805! C0805C180J5GACTU! Kemet! 80"C0805C180J5G! Mouser!
4! 1! D1! Diode,!Schottky!0.5A!20V! MBR0520LT1G! ONSemi! 863"MBR0520LT1G! Mouser!
5! 1! J1,J2! Headers,!36PS!1!Row! 68000"136HLF! FCI! 649"68000"136HLF! Mouser!
Connector,!Mini"B!Recept!
6! 1! J4! Rt.!Angle! 67503"1020! Molex! 538"67503"1020! Mouser!
7! 1! J5! Headers,!72PS!2!Rows! 67996"272HLF! FCI! 649"67996"272HLF! Mouser!
LED,!Super!Bright!RED!
100mcd!640nm!120degree!
8! 1! LD1! 0805! APT2012SRCPRV! Kingbright! 604"APT2012SRCPRV! Mouser!
LED,!Super!Bright!GREEN!
50mcd!570nm!110degree!
9! 1! LD2! 0805! APHCM2012CGCK"F01! Kingbright! 604"APHCM2012CGCK! Mouser!
LED,!Super!Bright!ORANGE!
160mcd!601nm!110degree!
10! 1! LD3! 0805! APHCM2012SECK"F01! Kingbright! 04"APHCM2012SECK! Mouser!
LED,!Super!Bright!BLUE!
80mcd!470nm!110degree!
11! 1! LD4! 0805! LTST"C170TBKT! Lite"On!Inc! 160"1579"1"ND! Digikey!
Resistor!Pack,!1K!+/"5%!
12! 1! R1! 62.5mW!4RES!SMD! YC164"JR"071KL! Yageo! YC164J"1.0KCT"ND! Digikey!
Resistor!Pack,!680!+/"5%!
13! 1! R2! 62.5mW!4RES!SMD! YC164"JR"07680RL! Yageo! YC164J"680CT"ND! Digikey!
Switch,!Momentary!Tact!
14! 1! SW1! SPST!150gf!3.0x2.5mm! B3U"1000P! Omron! SW1020CT"ND! Digikey!
IC,!Microcontroller!RISC!
16kB!Flash,!0.5kB!EEPROM,!
15! 1! U1! 23!I/O!Pins! ATmega168"20AU! Atmel! 556"ATMEGA168"20AU! Mouser!
IC,!USB!to!SERIAL!UART!28!
16! 1! U2! Pins!SSOP! FT232RL! FTDI! 895"FT232RL! Mouser!
IC,!Voltage!regulator!5V,!
17! 1! U3! 500mA!SOT"223! UA78M05CDCYRG3! TI! 595"UA78M05CDCYRG3! Mouser!
Cystal,!16MHz!+/"20ppm!
18! 1! Y1! HC"49/US!Low!Profile! ABL"16.000MHZ"B2! Abracon! 815"ABL"16"B2! Mouser!
HC Serial Bluetooth Products
User Instructional Manual
1 Introduction
HC serial Bluetooth products consist of Bluetooth serial interface module and Bluetooth adapter, such
as:
(1) Bluetooth serial interface module:
Industrial level: HC-03, HC-04(HC-04-M, HC-04-S)
Civil level: HC-05, HC-06(HC-06-M, HC-06-S)
HC-05-D, HC-06-D (with baseboard, for test and evaluation)
(2) Bluetooth adapter:
HC-M4
HC-M6
This document mainly introduces Bluetooth serial module. Bluetooth serial module is used for
converting serial port to Bluetooth. These modules have two modes: master and slaver device. The
device named after even number is defined to be master or slaver when out of factory and can’t be
changed to the other mode. But for the device named after odd number, users can set the work mode
(master or slaver) of the device by AT commands.
HC-04 specifically includes:
Master device: HC-04-M, M=master
Slave device: HC-04-S, S=slaver
The default situation of HC-04 is slave mode. If you need master mode, please state it clearly or
place an order for HC-O4-M directly.The naming rule of HC-06 is same.
When HC-03 and HC-05 are out of factory, one part of parameters are set for activating the device.
The work mode is not set, since user can set the mode of HC-03, HC-05 as they want.
The main function of Bluetooth serial module is replacing the serial port line, such as:
1. There are two MCUs want to communicate with each other. One connects to Bluetooth master
device while the other one connects to slave device. Their connection can be built once the pair is made.
This Bluetooth connection is equivalently liked to a serial port line connection including RXD, TXD
signals. And they can use the Bluetooth serial module to communicate with each other.
2. When MCU has Bluetooth salve module, it can communicate with Bluetooth adapter of
computers and smart phones. Then there is a virtual communicable serial port line between MCU and
computer or smart phone.
3. The Bluetooth devices in the market mostly are salve devices, such as Bluetooth printer,
Bluetooth GPS. So, we can use master module to make pair and communicate with them.
Bluetooth Serial module’s operation doesn’t need drive, and can communicate with the other
Bluetooth device who has the serial. But communication between two Bluetooth modules requires at
least two conditions:
(1) The communication must be between master and slave.
(2) The password must be correct.
However, the two conditions are not sufficient conditions. There are also some other conditions
basing on different device model. Detailed information is provided in the following chapters.
In the following chapters, we will repeatedly refer to Linvor’s (Formerly known as Guangzhou HC
Information Technology Co., Ltd.) material and photos.
2 Selection of the Module
The Bluetooth serial module named even number is compatible with each other; The salve module
is also compatible with each other. In other word, the function of HC-04 and HC-06, HC-03 and HC-05
are mutually compatible with each other. HC-04 and HC-06 are former version that user can’t reset the
work mode (master or slave). And only a few AT commands and functions can be used, like reset the
name of Bluetooth (only the slaver), reset the password, reset the baud rate and check the version
number. The command set of HC-03 and HC-05 are more flexible than HC-04 and HC-06’s. Generally,
the Bluetooth of HC-03/HC-05 is recommended for the user.
Here are the main factory parameters of HC-05 and HC-06. Pay attention to the differences:
HC-05 HC-06
Master and slave mode can be switched Master and slave mode can’t be switched
Bluetooth name: HC-05 Bluetooth name: linvor
Password:1234 Password:1234
Master role: have no function to remember the last
paired salve device. It can be made paired to any
slave device. In other words, just set Master role: have paired memory to remember
AT+CMODE=1 when out of factory. If you want last slave device and only make pair with that
HC-05 to remember the last paired slave device device unless KEY (PIN26) is triggered by high
address like HC-06, you can set AT+CMODE=0 level. The default connected PIN26 is low level.
after paired with the other device. Please refer the
command set of HC-05 for the details.
Pairing: The master device can not only make pair
with the specified Bluetooth address, like
cell-phone, computer adapter, slave device, but Pairing: Master device search and make pair with
also can search and make pair with the slave the slave device automatically.
device automatically. Typical method: On some specific conditions,
Typical method: On some specific conditions, master and slave device can make pair with each
master device and slave device can make pair with other automatically.
each other automatically. (This is the default
method.)
Multi-device communication: There is only point Multi-device communication: There is only point
to point communication for modules, but the to point communication for modules, but the
adapter can communicate with multi-modules. adapter can communicate with multi-modules.
AT Mode 1: After power on, it can enter the AT
mode by triggering PIN34 with high level. Then
the baud rate for setting AT command is equal to
the baud rate in communication, for example:
AT Mode: Before paired, it is at the AT mode.
9600.
After paired it’s at transparent communication.
AT mode 2: First set the PIN34 as high level, or
while on powering the module set the PIN34 to be
high level, the Baud rate used here is 38400 bps.
Notice: All AT commands can be operated only
when the PIN34 is at high level. Only part of the
AT commands can be used if PIN34 doesn’t keep
the high level after entering to the AT mode.
Through this kind of designing, set permissions for
the module is left to the user’s external control
circuit, that makes the application of HC-05 is very
flexible.
During the process of communication, the module
can enter to AT mode by setting PIN34 to be high
level. By releasing PIN34, the module can go back During the communication mode, the module
to communication mode in which user can inquire can’t enter to the AT mode.
some information dynamically. For example, to
inquire the pairing is finished or not.
Default communication baud rate: 9600, Default communication baud rate: 9600,
4800-1.3M are settable. 1200-1.3M are settable.
KEY: PIN34, for entering to the AT mode. KEY: PIN26, for master abandons memory.
LED1: PIN31, indicator of Bluetooth mode. Slow
flicker (1Hz) represents entering to the AT mode2,
while fast flicker(2Hz) represents entering to the LED: The flicker frequency of slave device is
AT mode1 or during the communication pairing. 102ms. If master device already has the memory
Double flicker per second represents pairing is of slave device, the flicker frequency during the
finished, the module is communicable. pairing is 110ms/s. If not, or master has emptied
LED2: PIN32, before pairing is at low level, after the memory, then the flicker frequency is 750m/s.
the pairing is at high level. After pairing, no matter it’s a master or slave
The using method of master and slaver’s indicator device, the LED PIN is at high level.
is the same. Notice: The LED PIN connects to LED+ PIN.
Notice: The PIN of LED1 and LED2 are connected
with LED+.
Consumption: During the pairing, the current is Consumption: During the pairing, the current is
fluctuant in the range of 30-40mA. The mean fluctuant in the range of 30-40 m. The mean
current is about 25mA. After paring, no matter current is about 25mA. After paring, no matter
processing communication or not, the current is processing communication or not, the current is
8mA. There is no sleep mode. This parameter is 8mA. There is no sleep mode. This parameter is
same for all the Bluetooth modules. same for all the Bluetooth modules.
Reset: PIN11, active if it’s input low level. It can Reset: PIN11, active if it’s input low level. It can
be suspended in using. be suspended in using.
Level: Civil Level: Civil
The table above that includes main parameters of two serial modules is a reference for user
selection.
HC-03/HC-05 serial product is recommended.
3. Information of Package
The PIN definitions of HC-03, HC-04, HC-05 and HC-06 are kind of different, but the package size
is the same: 28mm * 15mm * 2.35mm.
The following figure 1 is a picture of HC-06 and its main PINs. Figure 2 is a picture of HC-05 and
its main PINs. Figure 3 is a comparative picture with one coin. Figure 4 is their package size information.
When user designs the circuit, you can visit the website of Guangzhou HC Information Technology Co.,
Ltd. (www.wavesen.com) to download the package library of protle version.
Figure 1 HC-06 Figure 2 HC-05
Figure 3 Comparative picture with one coin
Figure 4 Package size information
4. The Using and Testing Method of HC-06 for the First Time
This chapter will introduce the using method of HC-06 in detail. User can test the module
according to this chapter when he or she uses the module at the first time.
PINs description:
PIN1 UART_TXD , TTL/CMOS level, UART Data output
PIN2 UART_RXD, TTL/COMS level, s UART Data input
RESET, the reset PIN of module, inputting low level can reset the module,
PIN11
when the module is in using, this PIN can connect to air.
VCC, voltage supply for logic, the standard voltage is 3.3V, and can work
PIN12
at 3.0-4.2V
PIN13 GND
PIN22 GND
LED, working mode indicator
Slave device: Before paired, this PIN outputs the period of 102ms square
wave. After paired, this PIN outputs high level.
Master device: On the condition of having no memory of pairing with a
PIN24
slave device, this PIN outputs the period of 110ms square wave. On the
condition of having the memory of pairing with a slave device, this PIN
outputs the period of 750ms square wave. After paired, this PIN outputs
high level.
For master device, this PIN is used for emptying information about
pairing. After emptying, master device will search slaver randomly, then
PIN26
remember the address of the new got slave device. In the next power on,
master device will only search this address.
(1) The circuit 1 (connect the module to 3.3V serial port of MCU) is showed by figure 5.
Figure 5 The circuit 1
In principle, HC-06 can work when UART_TXD, UART_RXD, VCC and GND are connected.
However, for better testing results, connecting LED and KEY are recommended (when testing the
master).
Where, the 3.3V TXD of MCU connects to HC-06’s UART_RXD, the 3.3V RXD of MCU connects
to HC-06’s UART_TXD, and 3.3V power and GND should be connected. Then the minimum system is
finished.
Note that, the PIN2:UART_RXD of Bluetooth module has no pull-up resistor. If the MCU TXD
doesn’t have pull-up function, then user should add a pull-up resistor to the UART_RXD. It may be easy
to be ignored.
If there are two MCU which connect to master and slave device respectively, then before
paired(LED will flicker) user can send AT commands by serial port when the system is power on. Please
refer to HC-04 and HC-06’s data sheet for detailed commands. In the last chapter, the command set will
be introduced. Please pay attention to that the command of HC-04/HC-06 doesn’t have terminator. For
example, consider the call command, sending out AT is already enough, need not add the CRLF
(carriage return line feed).
If the LED is constant lighting, it indicates the pairing is finished. The two MCUs can communicate
with each other by serial port. User can think there is a serial port line between two MCUs.
(2) The circuit 2 (connect the module to 5V serial port of MCU) is showed by figure 6.
Figure 6 is the block diagram of Bluetooth baseboard. This kind of circuit can amplify Bluetooth
module’s operating voltage to 3.1-6.5V. In this diagram, the J1 port can not only be connected with
MCU system of 3.3V and 5V, but also can be connected with computer serial port.
Figure 6 The circuit 2
(3) AT command test
Before paired, the mode of HC-04 and HC-06 are AT mode.
On the condition of 9600N81, OK will be received when user send the two letters AT. Please refer to the
last chapter of datasheet for other commands of HC-06. Please pay attention to that sending out AT is
already enough, need not add the CRLF (carriage return line feed).
The command set of Version V1.4 doesn’t include parity. The version V1.5 and its later version
have parity function. Moreover, there are three more commands of V1.5 than V1.4. They are:
No parity (default) AT+PN
Odd parity AT+PO
Even parity AT+PE
Do not let the sending frequency of AT command of HC-06 exceed 1Hz, because the command of
HC-06 end or not is determined by the time interval.
(4) Pairing with adapter
User can refer to the download center of the company’s website for “The Introduction of IVT” that
introduces the Bluetooth module makes pair with computer adapter. That document taking HC-06-D for
example introduces how the serial module makes pair with the adapter. That method is like to make pair
with cell-phone. But the difference is that cell-phone need a third-party communication software to help.
It’s liked the kind of PC serial helper of and the hyper terminal. A software named “PDA serial helper”
provided by our company is suitable for WM system. It has been proven that this serial module is
supported by many smart phone systems’ Bluetooth, such as, sybian, android, windows mobile and etc.
(5) Pairing introduction
HC-06 master device has no memory before the first use. If the password is correct, the mater
device will make pair with the slave device automatically in the first use. In the following use, the
master device will remember the Bluetooth address of the last paired device and search it. The searching
won’t stop until the device is found. If master device’s PIN26 is input high level, the device will lose the
memory. In that occasion, it’ll search the proper slave device like the first use. Based on this function,
the master device can be set to make pair with the specified address or any address by user.
(6) Reset new password introduction
User can set a new password for the HC-06 through AT+PINxxxx command. But the new password
will become active after discharged all the energy of the module. If the module still has any energy, the
old one is still active. In the test, for discharging all the system energy and activating the new password,
we can connect the power supply PIN with GND about 20 seconds after the power is cut off. Generally,
shutting down the device for 30 minutes also can discharge the energy, if there is no peripheral circuit
helps discharge energy. User should make the proper way according to the specific situation.
(7) Name introduction
If the device has no name, it’s better that user doesn’t try to change the master device name. The
name should be limited in 20 characters.
Summary: The character of HC-06: 1 not many command 2 easy for application 3 low price. It’s
good for some specific application. HC-04 is very similar with HC-06. Their only one difference is
HC-04 is for industry, HC-06 is for civil. Except this, they don’t have difference.
The following reference about HC-04 and HC-06 can be downloaded from company website
www.wavesen.com:
HC-06 datasheet .pdf (the command set introduction is included)
HC-04 datasheet .pdf (the command set introduction is included)
IVT BlueSoleil-2.6 (IVT Bluetooth drive test version)
Bluetooth FAQ.pdf
HC-04-D(HD-06-D)datasheet(English).pdf
HC-06-AT command software (test version) (some commands in V1.5 is not supported by V1.4)
PCB package of Bluetooth key modules (PCB package lib in protel)
IVT software manual.pdf (introduce how to operate the modern and make pair
with Bluetooth module)
PDA serial test helper.exe (serial helper used for WM system)
5 manual for the first use of HC-05
This chapter will introduce how to test and use the HC-05 if it’s the first time for user to operate it.
(1) PINs description
PIN1 UART_TXD, Bluetooth serial signal sending PIN, can connect with MCU’s RXD PIN
UART_RXD, Bluetooth serial signal receiving PIN, can connect with the MCU’s TXD PIN,
PIN2
there is no pull-up resistor in this PIN. But It needs to be added an eternal pull-up resistor.
RESET, the reset PIN of module, inputting low level can reset the module, when the module
PIN11
is in using, this PIN can connect to air.
PIN12 VCC, voltage supply for logic, the standard voltage is 3.3V, and can work at 3.0-4.2V
PIN13 GND
LED1, indicator of work mode. Has 3 modes:
When the module is supplied power and PIN34 is input high level, PIN31 output 1Hz square
wave to make the LED flicker slowly. It indicates that the module is at the AT mode, and the
baud rate is 38400;
When the module is supplied power and PIN34 is input low level, PIN31 output 2Hz square
wave to make the LED flicker quickly. It indicates the module is at the pairable mode. If
PIN31
PIN34 is input high level, then the module will enter to AT mode, but the output of PIN31 is
still 2Hz square wave.
After the pairing, PIN31 output 2Hz square ware.
Note: if PIN34 keep high level, all the commands in the AT command set can be in
application. Otherwise, if just excite PIN34 with high level but not keep, only some command
can be used. More information has provided at chapter 2.
Output terminal. Before paired, it output low level. Once the pair is finished, it output high
PIN32
level.
Mode switch input. If it is input low level, the module is at paired or communication mode. If
it’s input high level, the module will enter to AT mode. Even though the module is at
PIN34
communication, the module can enter to the AT mode if PIN34 is input high level. Then it will
go back to the communication mode if PIN34 is input low level again.
(2) Application circuit 1 (connect to the 3.3V system)
Figure 7 Application 1
(3) Application circuit 2 (connect to 5V serial system or PC serial)
Figure 8 Application circuit 2
(4) AT command test
This chapter introduces some common commands in use. The detail introduction about HC-05
command is in HC-0305 AT command set.
Enter to AT mode:
Way1: Supply power to module and input high level to PIN34 at the same time, the module will enter to
AT mode with the baud rate-38400.
Way2: In the first step, supply power to module; In the second step, input high level to PIN34. Then the
module will enter to AT mode with the baud rate-9600. Way1 is recommended.
Command structure: all command should end up with “\r\n” (Hex: 0X0D X0A) as the terminator. If
the serial helper is installed, user just need enter “ENTER” key at the end of command.
Reset the master-slave role command:
AT+ROLE=0 ----Set the module to be salve mode. The default mode is salve.
AT+ROLE=1 ----Set the module to be master mode.
Set memory command:
AT+CMODE=1
Set the module to make pair with the other random Bluetooth module (Not specified address). The
default is this mode.
AT+CMODE=1
Set the module to make pair with the other Bluetooth module (specified address). If set the module
to make pair with random one first, then set the module to make pair with the Bluetooth module has
specified address. Then the module will search the last paired module until the module is found.
Reset the password command
AT+PSWD=XXXX
Set the module pair password. The password must be 4-bits.
Reset the baud rate
AT+UART== ,,.
More information is provided at HC-0305 command set
Example:
AT+UART=9600,0,0 ----set the baud rate to be 9600N81
Reset the Bluetooth name
AT+NAME=XXXXX
Summary:
HC-05 has many functions and covers all functions of HC-06. The above commands are the most
common ones. Besides this, HC-05 leaves lots of space for user. So HC-05 is better than HC-06 and
recommended. HC-03 is similar with HC-05. The above introduction also suits HC-03
The following reference about HC-03 and HC-05 can be downloaded from company website
www.wavesen.com:
HC-03 datasheet .pdf (the command set introduction is included)
HC-05 datasheet .pdf (the command set introduction is included)
IVT BlueSoleil-2.6 (IVT Bluetooth drive test version)
Bluetooth FAQ.pdf
PCB package of Bluetooth key modules (PCB package lib in protel)
IVT software manual.pdf (introduce how to operate the modern and make pair with
Bluetooth module)
PDA serial test helper.exe (serial helper used for WM system)
HC-03/05 Bluetooth serial command set.pdf
6. Ordering information
The website of Guangzhou HC Information Technology Co., Ltd is www.wavesen.com The contact
information is provided at the company website.
Order Way: If you want our product, you can give order to the production center of our company
directly or order it in Taobao. There is a link to Taobao in our company website.
Package: 50 pieces chips in an anti-static blister package. The weight of a module is about 0.9g.
The weight of a package is about 50g.
Please provide the product’s model when you order:
HC-04-M HC-04 master module
HC-04-S HC-04 slave module
HC-06-M HC-06 master module
HC-06-S HC-06 slave module
HC-03
HC-05 HC-03/05 can be preset to be master module or slave module.
Product Sample & Technical Tools & Support &
Folder Buy Documents Software Community
LM2598
SNVS125D –MARCH 1998–REVISED MAY 2016
LM2598 SIMPLE SWITCHER® Power Converter 150-kHz 1-A Step-Down Voltage Regulator,
With Features
1 Features 3 Description
• 3.3-V, 5-V, 12-V, and Adjustable Output Versions The LM2598 series of regulators are monolithic1
• Adjustable Version Output Voltage Range, 1.2-V integrated circuits that provide all the active functionsfor a step-down (buck) switching regulator, capable of
to 37-V ±4% Max Over Line and Load Conditions driving a 1-A load with excellent line and load
• 1-A Output Current regulation. These devices are available in fixed output
• Available in 7-Pin TO-220 and DDPAK (Surface voltages of 3.3 V, 5 V, 12 V, and an adjustable output
Mount) Package version.
• Input Voltage Range Up to 40 V The LM2598 is a member of the LM259x family.
• Excellent Line and Load Regulation Specifications Requiring a minimum number of external
• 150 kHz Fixed Frequency Internal Oscillator components, these regulators are simple to use and
• Shutdown/Soft-start include internal frequency compensation, improvedline and load specifications, fixed-frequency oscillator,
• Out of Regulation Error Flag Shutdown/Soft-start, error flag delay and error flag
• Error Output Delay output.
• Low Power Standby Mode, IQ, Typically 85 μA The LM2598 series operates at a switching frequency
• High Efficiency of 150 kHz, thus allowing smaller sized filter
• Uses Readily Available Standard Inductors components than what would be required with lower-
frequency switching regulators. Available in a
• Thermal Shutdown and Current Limit Protection standard 7-lead TO-220 package with several
different lead bend options, and a 7-lead DDPAK
2 Applications surface mount package. Typically, for output voltages
• Simple High-Efficiency Step-down (Buck) less than 12 V, and ambient temperatures less than
Regulator 50°C, no heat sink is required.
• Efficient Preregulator for Linear Regulators Device Information(1)
• On-Card Switching Regulators PART NUMBER PACKAGE BODY SIZE (NOM)
• Positive to Negative Converter TO-220 (7) 14.986 mm × 10.16 mm
LM2598
TO-263 (7) 10.10 mm × 8.89 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application
Fixed output voltage versions
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM2598
SNVS125D –MARCH 1998–REVISED MAY 2016 www.ti.com
Table of Contents
1 Features .................................................................. 1 8.1 Overview ................................................................. 11
2 Applications ........................................................... 1 8.2 Functional Block Diagram ....................................... 11
3 Description ............................................................. 1 8.3 Feature Description................................................. 11
4 Revision History..................................................... 2 8.4 Device Functional Modes........................................ 16
5 Description (continued)......................................... 3 9 Application and Implementation ........................ 17
6 Pin Configuration and Functions ......................... 3 9.1 Application Information............................................ 17
7 Specifications 9.2 Typical Application .................................................. 28......................................................... 4
7.1 Absolute Maximum Ratings ..................................... 4 10 Power Supply Recommendations ..................... 37
7.2 ESD Ratings.............................................................. 4 11 Layout................................................................... 37
7.3 Recommended Operating Conditions....................... 4 11.1 Layout Guidelines ................................................. 37
7.4 Thermal Information .................................................. 4 11.2 Layout Examples................................................... 37
7.5 Electrical Characteristics – 3.3-V Version................. 5 11.3 Thermal Considerations ........................................ 38
7.6 Electrical Characteristics – 5-V Version.................... 5 12 Device and Documentation Support ................. 40
7.7 Electrical Characteristics – 12-V Version.................. 5 12.1 Community Resources.......................................... 40
7.8 Electrical Characteristics – Adjustable Voltage 12.2 Trademarks ........................................................... 40
Version ....................................................................... 6 12.3 Electrostatic Discharge Caution............................ 40
7.9 Electrical Characteristics – All Output Voltage 12.4 Glossary ................................................................ 40
Versions ..................................................................... 6 13 Mechanical, Packaging, and Orderable
7.10 Typical Characteristics ............................................ 8 Information ........................................................... 40
8 Detailed Description ............................................ 11
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (April 2013) to Revision D Page
• Added ESD Ratings table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section. ................................................................................................. 1
• Removed all references to design software Switchers Made Simple .................................................................................... 1
Changes from Revision B (April 2013) to Revision C Page
• Changed layout of National Semiconductor Data Sheet to TI format .................................................................................. 38
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5 Description (continued)
A standard series of inductors (both through hole and surface mount types) are available from several different
manufacturers optimized for use with the LM2598. This feature greatly simplifies the design of switch-mode
power supplies.
Other features include a specified ±4% tolerance on output voltage under all conditions of input voltage and
output load conditions, and ±15% on the oscillator frequency. External shutdown is included, featuring typically
85-μA standby current. Self-protection features include a two stage current limit for the output switch and an
overtemperature shutdown for complete protection under fault conditions.
6 Pin Configuration and Functions
NDZ Package
7-Pin TO-220 KTW Package
Top View 7-Pin TO-263
Top View
Pin Functions
PIN
I/O DESCRIPTION
NO. NAME
Internal switch. The voltage at this pin switches between approximately (+VIN – VSAT) and
1 Output O approximately –0.5 V, with a duty cycle of VOUT / VIN. To minimize coupling to sensitive
circuitry, the PCB copper area connected to this pin must be kept to a minimum.
This is the positive input supply for the IC switching regulator. A suitable input bypass
2 +VIN I capacitor must be present at this pin to minimize voltage transients and to supply the
switching currents required by the regulator.
Open collector output that provides a low signal (flag transistor ON) when the regulated output
3 Error Flag O voltage drops more than 5% from the nominal output voltage. On start up, Error Flag is lowuntil VOUT reaches 95% of the nominal output voltage and a delay time determined by the
Delay pin capacitor. This signal can be used as a reset to a microprocessor on power-up. (1)
4 Ground — Circuit ground.
At power-up, this pin can be used to provide a time delay between the time the regulated
5 Delay O output voltage reaches 95% of the nominal output voltage, and the time the error flag output
goes high. (1)
6 Feedback I Senses the regulated output voltage to complete the feedback loop.
This dual function pin provides the following features: (a) Allows the switching regulator circuit
7 Shutdown/Soft- to be shut down using logic level signals thus dropping the total input supply current tostart I approximately 80 μA. (b) Adding a capacitor to this pin provides a soft-start feature which
minimizes start-up current and provides a controlled ramp up of the output voltage. (1)
(1) If any of the above three features (Shutdown/Soft-start, Error Flag, or Delay) are not used, the respective pins must be left open.
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
MIN MAX UNIT
Maximum supply voltage, VIN 45 V
SD/SS pin input voltage (3) 6 V
Delay pin voltage (3) 1.5 V
Flag pin voltage –0.3 45 V
Feedback pin voltage –0.3 25 V
Output voltage to ground (steady state) –1 V
Power dissipation Internally limited
Vapor phase (60 s) 215
KTW package
Lead temperature Infrared (10 s) 245 °C
NDZ package (soldering, 10 s) 260
Maximum junction temperature 150 °C
Storage temperature, Tstg –65 150 °C
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) Voltage internally clamped. If clamp voltage is exceeded, limit current to a maximum of 1 mA.
7.2 ESD Ratings
VALUE UNIT
V(ESD) Electrostatic discharge Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1) (2) ±2000 V
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
(2) The human body model is a 100-pF capacitor discharged through a 1.5k resistor into each pin.
7.3 Recommended Operating Conditions
MIN MAX UNIT
Supply voltage 4.5 40 V
Temperature –25 125 °C
7.4 Thermal Information
LM2598
THERMAL METRIC (1) KTW (TO-263) NDZ (TO-220) UNIT
7 PINS 7 PINS
See (4) — 50
(5)
R Junction-to-ambient thermal resistance (2) (3)
See 50 —
θJA (6) °C/WSee 30 —
See (7) 20 —
RθJC(top) Junction-to-case (top) thermal resistance 2 2 °C/W
(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
(2) The package thermal impedance is calculated in accordance to JESD 51-7.
(3) Thermal Resistances were simulated on a 4 -layer, JEDEC board.
(4) Junction to ambient thermal resistance (no external heat sink) for the package mounted TO-220 package mounted vertically, with the
leads soldered to a printed circuit board with (1 oz.) copper area of approximately 1 in2.
(5) Junction to ambient thermal resistance with the TO-263 package tab soldered to a single sided printed circuit board with 0.5 in2 of (1
oz.) copper area.
(6) Junction to ambient thermal resistance with the TO-263 package tab soldered to a single sided printed circuit board with 2.5 in2 of (1
oz.) copper area.
(7) Junction to ambient thermal resistance with the TO-263 package tab soldered to a double sided printed circuit board with 3 in2 of (1 oz.)
copper area on the LM2598S side of the board, and approximately 16 in2 of copper on the other side of the PCB.
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7.5 Electrical Characteristics – 3.3-V Version
Specifications are for TJ = 25°C, unless otherwise specified.
PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) UNIT
SYSTEM PARAMETERS (3) (see Figure 42 and Figure 45 for test circuits)
TJ = 25°C 3.168 3.3 3.432
V Output voltage 4.75 V ≤ VIN ≤ 40 V,OUT 0.1 A ≤ I ≤ 1 A Over full operating VLOAD temperature range 3.135 3.465
η Efficiency VIN = 12 V, ILOAD = 1 A 78%
(1) All room temperature limits are 100% production tested. All limits at temperature extremes are specified via correlation using standard
Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2598 is used as shown in the Figure 42 and Figure 45, system performance is as shown in system parameters of Electrical
Characteristics.
7.6 Electrical Characteristics – 5-V Version
Specifications are for TJ = 25°C, unless otherwise specified.
PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) UNIT
SYSTEM PARAMETERS (3) (see Figure 42 and Figure 45 for test circuits)
TJ = 25°C 4.8 5 5.2
VOUT Output voltage
7 V ≤ VIN ≤ 40 V,
0.1 A ≤ I ≤ 1 A Over full operating VLOAD temperature range 4.75 5.25
η Efficiency VIN = 12 V, ILOAD = 1 A 82%
(1) All room temperature limits are 100% production tested. All limits at temperature extremes are specified via correlation using standard
Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2598 is used as shown in the Figure 42 and Figure 45, system performance is as shown in system parameters of Electrical
Characteristics.
7.7 Electrical Characteristics – 12-V Version
Specifications are for TJ = 25°C, unless otherwise specified.
PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) UNIT
SYSTEM PARAMETERS (3) (see Figure 42 and Figure 45 for test circuits)
T = 25°C 11.52 12 12.48
V Output voltage 15 V ≤ VIN ≤ 40 V,
J
OUT 0.1 A ≤ I Over full operating VLOAD ≤ 1 A temperature range 11.4 12.6
η Efficiency VIN = 25 V, ILOAD = 1 A 90%
(1) All room temperature limits are 100% production tested. All limits at temperature extremes are specified via correlation using standard
Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2598 is used as shown in the Figure 42 and Figure 45, system performance is as shown in system parameters of Electrical
Characteristics.
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7.8 Electrical Characteristics – Adjustable Voltage Version
Specifications are for TJ = 25°C, unless otherwise specified.
PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) UNIT
SYSTEM PARAMETERS (3) (see Figure 42 and Figure 45 for test circuits)
4.5 V ≤ VIN ≤ 40 V, 0.1 A ≤ ILOAD ≤ 1 A 1.23
VFB Feedback voltage VOUT programmed for 3 V,
TJ = 25°C 1.193 1.267 V
circuit of Figure 42 and Over full operating
Figure 45 temperature range 1.18 1.28
η Efficiency VIN = 12 V, VOUT = 3 V, ILOAD = 1 A 78%
(1) All room temperature limits are 100% production tested. All limits at temperature extremes are specified via correlation using standard
Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance.
When the LM2598 is used as shown in the Figure 42 and Figure 45, system performance is as shown in system parameters of Electrical
Characteristics.
7.9 Electrical Characteristics – All Output Voltage Versions
Specifications are for TJ = 25°C unless otherwise noted. Unless otherwise specified, VIN = 12 V for the 3.3-V, 5-V, and
Adjustable version and VIN = 24 V for the 12-V version. ILOAD = 500 mA
PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) UNIT
DEVICE PARAMETERS
T = 25°C 10 50
I Feedback bias Adjustable version only,
J
b current V = 1.3 V Over full operating nAFB temperature range 100
TJ = 25°C 127 150 173
f (3)O Oscillator frequency See Over full operating kHz
temperature range 110 173
TJ = 25°C 1 1.2
VSAT Saturation voltage IOUT = 1 A (4) (5) Over full operating V
temperature range 1.3
Max duty cycle (ON) See (5) 100%
DC Minimum duty cycle (6)
(OFF) See 0%
TJ = 25°C 1.2 1.5 2.4
ICL Current limit Peak current (4) (5) Over full operating A
temperature range 1.15 2.6
Output leakage Output = 0 V, see (4) (6) (7) 50 μAIL current Output = –1 V 2 15 mA
I Operating quiescentQ current SD/SS pin open
(6) 5 10 mA
TJ = 25°C 85 200
I Current standbySTBY quiescent SD/SS pin = 0 V
(7) Over full operating μA
temperature range 250
(1) All room temperature limits are 100% production tested. All limits at temperature extremes are specified via correlation using standard
Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Typical numbers are at 25°C and represent the most likely norm.
(3) The switching frequency is reduced when the second stage current limit is activated. The amount of reduction is determined by the
severity of current overload.
(4) No diode, inductor or capacitor connected to output pin.
(5) Feedback pin removed from output and connected to 0 V to force the output transistor switch ON.
(6) Feedback pin removed from output and connected to 12 V for the 3.3-V, 5-V, and the Adjustable version, and 15 V for the 12-V version,
to force the output transistor switch OFF.
(7) VIN = 40 V.
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Electrical Characteristics – All Output Voltage Versions (continued)
Specifications are for TJ = 25°C unless otherwise noted. Unless otherwise specified, VIN = 12 V for the 3.3-V, 5-V, and
Adjustable version and VIN = 24 V for the 12-V version. ILOAD = 500 mA
PARAMETER TEST CONDITIONS MIN (1) TYP (2) MAX (1) UNIT
SHUTDOWN AND SOFT-START CONTROL (see Figure 42 and Figure 45 for test circuits)
TJ = 25°C 1.3
Shutdown threshold Low, (Shutdown Mode), over full operating temperatureVSD voltage range
0.6 V
High, (Soft-start Mode), over full operating temperature
range 2
VOUT = 20% of nominal output voltage 2VSS Soft-start voltage VVOUT = 100% of nominal output voltage 3
ISD Shutdown current VSHUTDOWN = 0.5 V 5 10 μA
ISS Soft-start current VSoft-start = 2.5 V 1.6 5 μA
FLAG AND DELAY CONTROL (see Figure 42 and Figure 45 for test circuits)
Regulator dropout
detector threshold Low (Flag ON) 92% 96% 98%
voltage
ISINK = 3 mA 0.3 V
VF Voltage flag output TJ = 25°C 0.7SAT saturation VDELAY = 0.5 V Over full operating V
temperature range 1
IF Flag output leakageL current VFLAG = 40 V 0.3 μA
1.25 V
Voltage delay pin
threshold Low (Flag ON) 1.21 V
High (Flag OFF) and VOUT Regulated 1.29
Delay pin source
current VDELAY = 0.5 V 3 6 μA
TJ = 25°C 55 350
Delay pin saturation Low (Flag ON) Over full operating mV
temperature range 400
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7.10 Typical Characteristics
Circuit of Figure 45
Figure 1. Normalized Output Voltage Figure 2. Line Regulation
Figure 3. Efficiency Figure 4. Switch Saturation Voltage
Figure 5. Switch Current Limit Figure 6. Dropout Voltage
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Typical Characteristics (continued)
Circuit of Figure 45
Figure 7. Operating Quiescent Current Figure 8. Shutdown Quiescent Current
Figure 9. Minimum Operating Supply Voltage Figure 10. Feedback Pin Bias Current
Figure 11. Flag Saturation Voltage Figure 12. Switching Frequency
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Typical Characteristics (continued)
Circuit of Figure 45
Figure 13. Soft-start Figure 14. Shutdown/Soft-start Current
Figure 15. Delay Pin Current Figure 16. Soft-start Response
Figure 17. Shutdown and Soft-start Threshold Voltage
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8 Detailed Description
8.1 Overview
The LM2598 SIMPLE SWITCHER® regulator is an easy-to-use, nonsynchronous, step-down DC-DC converter
with a wide input voltage range up to 40 V. The regulator is capable of delivering up to 1-A DC load current with
excellent line and load regulation. These devices are available in fixed output voltages of 3.3-V, 5-V, 12-V and an
adjustable output version. The family requires few external components, and the pin arrangement was designed
for simple, optimum PCB layout.
8.2 Functional Block Diagram
8.3 Feature Description
8.3.1 SHUTDOWN and Soft-Start
The circuit shown in Figure 20 is a standard buck regulator with 24-VIN, 12-VOUT, 280-mA load, and using a
0.068-μF soft-start capacitor. The photo in Figure 18 and Figure 19 show the effects of Soft-start on the output
voltage, the input current, with, and without a soft-start capacitor. Figure 18 also shows the error flag output
going high when the output voltage reaches 95% of the nominal output voltage. The reduced input current
required at start-up is very evident when comparing the two photos. The Soft-start feature reduces the start-up
current from 1 A down to 240 mA, and delays and slows down the output voltage rise time.
This reduction in start-up current is useful in situations where the input power source is limited in the amount of
current it can deliver. In some applications Soft-start can be used to replace undervoltage lockout or delayed
start-up functions.
If a very slow output voltage ramp is desired, the Soft-start capacitor can be made much larger. Many seconds or
even minutes are possible.
If only the shutdown feature is required, the Soft-start capacitor can be eliminated.
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Feature Description (continued)
Figure 18. Output Voltage, Input Current, and Error Flag
Signal at Start-Up With Soft-start
Figure 19. Output Voltage and Input Current at Start-Up
Without Soft-start
Figure 20. Typical Circuit Using Shutdown/Soft-start and Error Flag Features
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Feature Description (continued)
Figure 21. Inverting –5-V Regulator With Shutdown and Soft-start
8.3.2 Inverting Regulator
The circuit in Figure 21 converts a positive input voltage to a negative output voltage with a common ground. The
circuit operates by bootstrapping the regulators ground pin to the negative output voltage, then grounding the
feedback pin, the regulator senses the inverted output voltage and regulates it.
This example uses the LM2598-5 to generate a –5-V output, but other output voltages are possible by selecting
other output voltage versions, including the adjustable version. Because this regulator topology can produce an
output voltage that is either greater than or less than the input voltage, the maximum output current greatly
depends on both the input and output voltage. The curve shown in Figure 22 provides a guide as to the amount
of output load current possible for the different input and output voltage conditions.
The maximum voltage appearing across the regulator is the absolute sum of the input and output voltage, and
this must be limited to a maximum of 40 V. In this example, when converting 20 V to –5 V, the regulator would
see 25 V between the input pin and ground pin. The LM2598 has a maximum input voltage rating of 40 V.
Figure 22. Maximum Load Current for Inverting
Regulator Circuit
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Feature Description (continued)
An additional diode is required in this regulator configuration. Diode D1 is used to isolate input voltage ripple or
noise from coupling through the CIN capacitor to the output, under light or no load conditions. Also, this diode
isolation changes the topology to closely resemble a buck configuration thus providing good closed loop stability.
A Schottky diode is recommended for low input voltages, (because of its lower voltage drop) but for higher input
voltages, a 1N5400 diode could be used.
Because of differences in the operation of the inverting regulator, the standard design procedure is not used to
select the inductor value. In the majority of designs, a 68-μH, 1.5-A inductor is the best choice. Capacitor
selection can also be narrowed down to just a few values. Using the values shown in Figure 21 provides good
results in the majority of inverting designs.
This type of inverting regulator can require relatively large amounts of input current when starting up, even with
light loads. Input currents as high as the LM2598 current limit (approximately 1.5 A) are required for 2 ms or
more, until the output reaches its nominal output voltage. The actual time depends on the output voltage and the
size of the output capacitor. Input power sources that are current limited or sources that can not deliver these
currents without getting loaded down, may not work correctly. Because of the relatively high start-up currents
required by the inverting topology, the soft-start feature shown in Figure 21 is recommended.
Also shown in Figure 21 are several shutdown methods for the inverting configuration. With the inverting
configuration, some level shifting is required, because the ground pin of the regulator is no longer at ground, but
is now at the negative output voltage. The shutdown methods shown accept ground referenced shutdown
signals.
8.3.3 Undervoltage Lockout
Some applications require the regulator to remain off until the input voltage reaches a predetermined voltage.
Figure 23 shows an undervoltage lockout feature applied to a buck regulator, while Figure 24 and Figure 25 are
for the inverting types (only the circuitry pertaining to the undervoltage lockout is shown). Figure 23 uses a Zener
diode to establish the threshold voltage when the switcher begins operating. When the input voltage is less than
the Zener voltage, resistors R1 and R2 hold the Shutdown or Soft-start pin low, keeping the regulator in the
shutdown mode. As the input voltage exceeds the Zener voltage, the Zener conducts, pulling the Shutdown/Soft-
start pin high, allowing the regulator to begin switching. The threshold voltage for the undervoltage lockout
feature is approximately 1.5 V greater than the Zener voltage.
Figure 23. Undervoltage Lockout for a Buck Regulator
Figure 24 and Figure 25 apply the same feature to an inverting circuit. Figure 24 features a constant threshold
voltage for turnon and turnoff (Zener voltage plus approximately 1 V). Because the SD/SS pin has an internal 7-V
zener clamp, R2 is required to limit the current into this pin to approximately 1 mA when Q1 is on. If hysteresis is
required, the circuit in Figure 25 has a turnon voltage which is different than the turnoff voltage. The amount of
hysteresis is approximately equal to the value of the output voltage.
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Feature Description (continued)
Figure 24. Undervoltage Lockout Without
Hysteresis for an Inverting Regulator
Figure 25. Undervoltage Lockout With
Hysteresis for an Inverting Regulator
8.3.4 Negative Voltage Charge Pump
Occasionally a low current negative voltage is required for biasing parts of a circuit. A simple method of
generating a negative voltage using a charge pump technique and the switching waveform present at the OUT
pin, is shown in Figure 26. This unregulated negative voltage is approximately equal to the positive input voltage
(minus a few volts), and can supply up to a 200 mA of output current. There is a requirement however, that there
be a minimum load of several hundred mA on the regulated positive output for the charge pump to work
correctly. Also, resistor R1 is required to limit the charging current of C1 to some value less than the LM2598
current limit (typically 1.5 A).
This method of generating a negative output voltage without an additional inductor can be used with other
members of the Simple Switcher Family, using either the buck or boost topology.
Figure 26. Charge Pump for Generating a
Low Current, Negative Output Voltage
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8.4 Device Functional Modes
8.4.1 Discontinuous Mode Operation
The selection guide chooses inductor values suitable for continuous mode operation, but for low current
applications or high input voltages, a discontinuous mode design may be a better choice. Discontinuous mode
would use an inductor that would be physically smaller, and would require only one half to one third the
inductance value required for a continuous mode design. The peak switch and inductor currents is higher in a
discontinuous design, but at these low load currents (200 mA and below), the maximum switch current is still less
than the switch current limit.
Discontinuous operation can have voltage waveforms that are considerably different than a continuous design.
The output pin (switch) waveform can have some damped sinusoidal ringing present (see Figure 46) This ringing
is normal for discontinuous operation, and is not caused by feedback loop instabilities. In discontinuous
operation, there is a period of time where neither the switch nor the diode are conducting, and the inductor
current has dropped to zero. During this time, a small amount of energy can circulate between the inductor and
the switch or diode parasitic capacitance causing this characteristic ringing. Normally this ringing is not a
problem, unless the amplitude becomes great enough to exceed the input voltage, and even then, there is very
little energy present to cause damage.
Different inductor types or core materials produce different amounts of this characteristic ringing. Ferrite core
inductors have very little core loss and therefore produce the most ringing. The higher core loss of powdered iron
inductors produce less ringing. If desired, a series RC could be placed in parallel with the inductor to dampen the
ringing.
Figure 27. Post Ripple Filter Waveform
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
9.1.1 Soft-Start Capacitor (CSS)
A capacitor on this pin provides the regulator with a Soft-start feature (slow start-up). When the DC input voltage
is first applied to the regulator, or when the Shutdown/Soft-start pin is allowed to go high, a constant current
(approximately 5 μA begins charging this capacitor). As the capacitor voltage rises, the regulator goes through
four operating regions (See the bottom curve in Figure 28).
1. Regulator in shutdown: When the SD/SS pin voltage is between 0 V and 1.3 V, the regulator is in shutdown,
the output voltage is zero, and the IC quiescent current is approximately 85 μA.
2. Regulator ON, but the output voltage is zero: With the SD/SS pin voltage between approximately 1.3 V and
1.8 V, the internal regulator circuitry is operating, the quiescent current rises to approximately 5 mA, but the
output voltage is still zero. Also, as the 1.3-V threshold is exceeded, the Soft-start capacitor charging current
decreases from 5 μA down to approximately 1.6 μA. This decreases the slope of capacitor voltage ramp.
3. Soft-start region: When the SD/SS pin voltage is between 1.8 V and 2.8 V at 25°C, the regulator is in a Soft-
start condition. The switch (Pin 1) duty cycle initially starts out very low, with narrow pulses and gradually get
wider as the capacitor SD/SS pin ramps up towards 2.8 V. As the duty cycle increases, the output voltage
also increases at a controlled ramp up. See the center curve in Figure 28. The input supply current
requirement also starts out at a low level for the narrow pulses and ramp up in a controlled manner. This is a
very useful feature in some switcher topologies that require large start-up currents (such as the inverting
configuration) which can load down the input power supply.
Note: The lower curve shown in Figure 28 shows the Soft-start region from 0% to 100%. This is not the duty
cycle percentage, but the output voltage percentage. Also, the Soft-start voltage range has a negative
temperature coefficient associated with it.
4. Normal operation: Above 2.8 V, the circuit operates as a standard pulse width modulated switching regulator.
The capacitor continues to charge up until it reaches the internal clamp voltage of approximately 7 V. If this
pin is driven from a voltage source, the current must be limited to about 1 mA.
If the part is operated with an input voltage at or below the internal soft-start clamp voltage of approximately 7 V,
the voltage on the SD/SS pin tracks the input voltage and can be disturbed by a step in the voltage. To maintain
proper function under these conditions, it is strongly recommended that the SD/SS pin be clamped externally
between the 3-V maximum soft-start threshold and the 4.5-V minimum input voltage. Figure 30 is an example of
an external approximately 3.7-V clamp that prevents a line-step related glitch but does not interfere with the soft-
start behavior of the device.
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Application Information (continued)
Figure 28. Soft-start, Delay, Error, Output
Figure 29. Timing Diagram for 5-V Output
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Application Information (continued)
VIN
LM2598
5 Q1
SD/SS
CSS Z1
3V
Figure 30. External 3.7-V Soft-Start Clamp
9.1.2 Delay Capacitor (CDELAY)
Provides delay for the error flag output. See the upper curve in Figure 28, and also refer to timing diagrams in
Figure 29. A capacitor on this pin provides a time delay between the time the regulated output voltage (when it is
increasing in value) reaches 95% of the nominal output voltage, and the time the error flag output goes high. A 3-
μA constant current from the delay pin charges the delay capacitor resulting in a voltage ramp. When this voltage
reaches a threshold of approximately 1.3 V, the open collector error flag output (or power OK) goes high. This
signal can be used to indicate that the regulated output has reached the correct voltage and has stabilized.
If, for any reason, the regulated output voltage drops by 5% or more, the error output flag (Pin 3) immediately
goes low (internal transistor turns on). The delay capacitor provides very little delay if the regulated output is
dropping out of regulation. The delay time for an output that is decreasing is approximately a 1000 times less
than the delay for the rising output. For a 0.1-μF delay capacitor, the delay time would be approximately 50 ms
when the output is rising and passes through the 95% threshold, but the delay for the output dropping would only
be approximately 50 μs.
The error flag output, RPull Up (or power OK), is the collector of a NPN transistor, with the emitter internally
grounded. To use the error flag, a pullup resistor to a positive voltage is required. The error flag transistor is
rated up to a maximum of 45 V and can sink approximately 3 mA. If the error flag is not used, it can be left open.
9.1.3 Feedforward Capacitor (CFF)
NOTE
Adjustable output voltage version only
Figure 45 shows a feedfoward capacitor across R2 which is used when the output voltage is greater than 10 V or
then COUT has a very low ESR. This capacitor adds lead compensation to the feedback loop and increases the
phase margin for better loop stability.
If the output ripple is large (> 5% of the nominal output voltage), this ripple can be coupled to the feedback pin
through the feedforward capacitor and cause the error comparator to trigger the error flag. In this situation,
adding a resistor, RFF, in series with the feedforward capacitor, approximately 3 times R1, attenuates the ripple
voltage at the feedback pin.
9.1.4 Input Capacitor (CIN)
A low ESR aluminum or tantalum bypass capacitor is required between the input pin and ground pin. The
capacitor must be located near the regulator using short leads. This capacitor prevents large voltage transients
from appearing at the input, and provides the instantaneous current required each time the switch turns on.
The important parameters for the Input capacitor are the voltage rating and the RMS current rating. Because of
the relatively high RMS currents flowing in a buck regulator's input capacitor, this capacitor must be chosen for
its RMS current rating rather than its capacitance or voltage ratings, although the capacitance value and voltage
rating are directly related to the RMS current rating.
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Application Information (continued)
The RMS current rating of a capacitor could be viewed as a capacitor's power rating. The RMS current flowing
through the capacitors internal ESR produces power which causes the internal temperature of the capacitor to
rise. The RMS current rating of a capacitor is determined by the amount of current required to raise the internal
temperature approximately 10°C above an ambient temperature of 105°C. The ability of the capacitor to dissipate
this heat to the surrounding air determines the amount of current the capacitor can safely sustain. Capacitors that
are physically large and have a large surface area typically has higher RMS current ratings. For a given capacitor
value, a higher voltage electrolytic capacitor is physically larger than a lower voltage capacitor, and thus be able
to dissipate more heat to the surrounding air, and therefore has a higher RMS current rating.
Figure 31. RMS Current Ratings for Low
ESR Electrolytic Capacitors (Typical)
Figure 32. Capacitor ESR vs Capacitor Voltage Rating
(Typical Low ESR Electrolytic Capacitor)
The consequences of operating an electrolytic capacitor above the RMS current rating is a shortened operating
life. The higher temperature speeds up the evaporation of the capacitor's electrolyte, resulting in eventual failure.
Selecting an input capacitor requires consulting the manufacturers data sheet for maximum allowable RMS ripple
current. For a maximum ambient temperature of 40°C, a general guideline would be to select a capacitor with a
ripple current rating of approximately 50% of the DC load current. For ambient temperatures up to 70°C, a
current rating of 75% of the DC load current would be a good choice for a conservative design. The capacitor
voltage rating must be at least 1.25 times greater than the maximum input voltage, and often a much higher
voltage capacitor is required to satisfy the RMS current requirements.
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Application Information (continued)
Figure 31 shows the relationship between an electrolytic capacitor value, its voltage rating, and the RMS current
it is rated for. These curves were obtained from the Nichicon PL series of low-ESR, high-reliability electrolytic
capacitors designed for switching regulator applications. Other capacitor manufacturers offer similar types of
capacitors, but always check the capacitor data sheet.
Standard electrolytic capacitors typically have much higher ESR numbers, lower RMS current ratings and
typically have a shorter operating lifetime.
Because of their small size and excellent performance, surface mount solid tantalum capacitors are often used
for input bypassing, but several precautions must be observed. A small percentage of solid tantalum capacitors
can short if the inrush current rating is exceeded. This can happen at turnon when the input voltage is suddenly
applied, and of course, higher input voltages produce higher inrush currents. Several capacitor manufacturers do
a 100% surge current testing on their products to minimize this potential problem. If high turnon currents are
expected, it may be necessary to limit this current by adding either some resistance or inductance before the
tantalum capacitor, or select a higher voltage capacitor. As with aluminum electrolytic capacitors, the RMS ripple
current rating must be sized to the load current.
9.1.5 Output Capacitor (COUT)
An output capacitor is required to filter the output and provide regulator loop stability. Low impedance or low ESR
Electrolytic or solid tantalum capacitors designed for switching regulator applications must be used. When
selecting an output capacitor, the important capacitor parameters are; the 100-kHz Equivalent Series Resistance
(ESR), the RMS ripple current rating, voltage rating, and capacitance value. For the output capacitor, the ESR
value is the most important parameter.
The output capacitor requires an ESR value that has an upper and lower limit. For low output ripple voltage, a
low ESR value is required. This value is determined by the maximum allowable output ripple voltage, typically 1%
to 2% of the output voltage. But if the selected capacitor's ESR is extremely low, there is a possibility of an
unstable feedback loop, resulting in an oscillation at the output. Using the capacitors listed in the tables, or
similar types, provides design solutions under all conditions.
If very low output ripple voltage (less than 15 mV) is required, see Output Voltage Ripple and Transients for a
post ripple filter.
An aluminum electrolytic capacitor's ESR value is related to the capacitance value and its voltage rating. In most
cases, higher voltage electrolytic capacitors have lower ESR values (see Figure 32). Often, capacitors with much
higher voltage ratings may be required to provide the low ESR values required for low output ripple voltage.
The output capacitor for many different switcher designs often can be satisfied with only three or four different
capacitor values and several different voltage ratings. See Figure 38 and Table 1 for typical capacitor values,
voltage ratings, and manufacturers capacitor types.
Electrolytic capacitors are not recommended for temperatures below –25°C. The ESR rises dramatically at cold
temperatures and typically rises 3X at –25°C and as much as 10X at –40°C. See curve shown in Figure 33.
Solid tantalum capacitors have a much better ESR specifications for cold temperatures and are recommended
for temperatures below –25°C.
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Application Information (continued)
Table 1. Output Capacitor and Feedforward Capacitor Selection Table
THROUGH-HOLE ELECTROLYTIC SURFACE-MOUNT TANTALUM
OUTPUT
VOLTAGE PANASONIC NICHICON PL
HFQ SERIES SERIES FEEDFORWARD
AVX TPS SPRAGUE
SERIES 595D SERIES FEEDFORWARD(V)
(μF/V) (μF/V) CAPACITOR (μF/V) (μF/V) CAPACITOR
1.2 330/50 330/50 0 330/6.3 330/6.3 0
4 220/25 220/25 4.7 nF 220/10 220/10 4.7 nF
6 220/25 220/25 3.3 nF 220/10 220/10 3.3 nF
9 180/25 180/25 1.5 nF 100/16 180/16 1.5 nF
12 120/25 120/25 1.5 nF 68/20 120/20 1.5 nF
15 120/25 120/25 1.5 nF 68/20 100/20 1.5 nF
24 82/35 82/35 1 nF 33/25 33/35 220 pF
28 82/50 82/50 1 nF 10/35 33/35 220 pF
9.1.6 Catch Diode
Buck regulators require a diode to provide a return path for the inductor current when the switch turns off. This
must be a fast diode and must be located close to the LM2598 using short leads and short printed circuit traces.
Because of their very fast switching speed and low forward voltage drop, Schottky diodes provide the best
performance, especially in low output voltage applications (5 V and lower). Ultra-fast recovery, or high-efficiency
rectifiers are also a good choice, but some types with an abrupt turnoff characteristic may cause instability or
EMI problems. Ultra-fast recovery diodes typically have reverse recovery times of 50 ns or less. Rectifiers such
as the 1N5400 series are much too slow and must not be used.
Figure 33. Capacitor ESR Change vs Temperature
9.1.7 Inductor Selection
All switching regulators have two basic modes of operation; continuous and discontinuous. The difference
between the two types relates to the inductor current, whether it is flowing continuously, or if it drops to zero for a
period of time in the normal switching cycle. Each mode has distinctively different operating characteristics,
which can affect the regulators performance and requirements. Most switcher designs operate in the
discontinuous mode when the load current is low.
The LM2598 (or any of the Simple Switcher family) can be used for both continuous or discontinuous modes of
operation.
In many cases the preferred mode of operation is the continuous mode. This mode offers greater output power,
lower peak switch, inductor and diode currents, and can have lower output ripple voltage. However, the
continuous mode requires larger inductor values to keep the inductor current flowing continuously, especially at
low output load currents or high input voltages.
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To simplify the inductor selection process, an inductor selection guide (nomograph) was designed (see Table 1
through Figure 37). This guide assumes that the regulator is operating in the continuous mode, and selects an
inductor that allows a peak-to-peak inductor ripple current to be a certain percentage of the maximum design
load current. This peak-to-peak inductor ripple current percentage is not fixed, but is allowed to change as
different design load currents are selected. (See Figure 34.)
Figure 34. (ΔIIND) Peak-to-Peak Inductor
Ripple Current (as a Percentage of the
Load Current) vs Load Current
By allowing the percentage of inductor ripple current to increase for low load currents, the inductor value and size
can be kept relatively low.
When operating in the continuous mode, the inductor current waveform ranges from a triangular to a sawtooth
type of waveform (depending on the input voltage), with the average value of this current waveform equal to the
DC output load current.
Inductors are available in different styles such as pot core, toroid, E-core, bobbin core, and so forth, as well as
different core materials, such as ferrites and powdered iron. The least expensive, the bobbin, rod or stick core,
consists of wire wound on a ferrite bobbin. This type of construction makes for an inexpensive inductor; however,
because the magnetic flux is not completely contained within the core, it generates more Electro-Magnetic
Interference (EMl). This magnetic flux can induce voltages into nearby printed circuit traces, thus causing
problems with both the switching regulator operation and nearby sensitive circuitry, and can give incorrect scope
readings because of induced voltages in the scope probe. Also seeOpen Core Inductors.
When multiple switching regulators are located on the same PCB, open core magnetics can cause interference
between two or more of the regulator circuits, especially at high currents. A torroid or E-core inductor (closed
magnetic structure) must be used in these situations.
The inductors listed in the selection chart include ferrite E-core construction for Schott, ferrite bobbin core for
Renco and Coilcraft, and powdered iron toroid for Pulse Engineering.
Exceeding an inductor's maximum current rating may cause the inductor to overheat because of the copper wire
losses, or the core may saturate. If the inductor begins to saturate, the inductance decreases rapidly and the
inductor begins to look mainly resistive (the DC resistance of the winding). This can cause the switch current to
rise very rapidly and force the switch into a cycle-by-cycle current limit, thus reducing the DC output load current.
This can also result in overheating of the inductor or the LM2598. Different inductor types have different
saturation characteristics, and this must be kept in mind when selecting an inductor.
The inductor manufacturer's data sheets include current and energy limits to avoid inductor saturation.
For continuous mode operation, see the inductor selection graphs in Figure 35 through Figure 38.
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Figure 35. LM2598-3.3 Figure 36. LM2598-5.0
Figure 37. LM2598-12 Figure 38. LM2598-ADJ
Table 2. Inductor Manufacturers Part Numbers
SCHOTTKY RENCO PULSE ENGINEERING COILCRAFT
INDUCTANCE CURRENT
(μH) (A) THROUGH SURFACE THROUGH SURFACE THROUGH SURFACE SURFACE
HOLE MOUNT HOLE MOUNT HOLE MOUNT MOUNT
L4 68 0.32 67143940 67144310 RL-1284-68-43 RL1500-68 PE-53804 PE-53804-S DO1608-68
L5 47 0.37 67148310 67148420 RL-1284-47-43 RL1500-47 PE-53805 PE-53805-S DO1608-473
L6 33 0.44 67148320 67148430 RL-1284-33-43 RL1500-33 PE-53806 PE-53806-S DO1608-333
L9 220 0.32 67143960 67144330 RL-5470-3 RL1500-220 PE-53809 PE-53809-S DO3308-224
L10 150 0.39 67143970 67144340 RL-5470-4 RL1500-150 PE-53810 PE-53810-S DO3308-154
L11 100 0.48 67143980 67144350 RL-5470-5 RL1500-100 PE-53811 PE-53811-S DO3308-104
L12 68 0.58 67143990 67144360 RL-5470-6 RL1500-68 PE-53812 PE-53812-S DO3308-683
L13 47 0.7 67144000 67144380 RL-5470-7 RL1500-47 PE-53813 PE-53813-S DO3308-473
L14 33 0.83 67148340 67148450 RL-1284-33-43 RL1500-33 PE-53814 PE-53814-S DO3308-333
L15 22 0.99 67148350 67148460 RL-1284-22-43 RL1500-22 PE-53815 PE-53815-S DO3308-223
L16 15 1.24 67148360 67148470 RL-1284-15-43 RL1500-15 PE-53816 PE-53816-S DO3308-153
L17 330 0.42 67144030 67144410 RL-5471-1 RL1500-330 PE-53817 PE-53817-S DO3316-334
L18 220 0.55 67144040 67144420 RL-5471-2 RL1500-220 PE-53818 PE-53818-S DO3316-224
L19 150 0.66 67144050 67144430 RL-5471-3 RL1500-150 PE-53819 PE-53819-S DO3316-154
L20 100 0.82 67144060 67144440 RL-5471-4 RL1500-100 PE-53820 PE-53820-S DO3316-104
L21 68 0.99 67144070 67144450 RL-5471-5 RL1500-68 PE-53821 PE-53821-S DO3316-683
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Table 2. Inductor Manufacturers Part Numbers (continued)
SCHOTTKY RENCO PULSE ENGINEERING COILCRAFT
INDUCTANCE CURRENT
(μH) (A) THROUGH SURFACE THROUGH SURFACE THROUGH SURFACE SURFACE
HOLE MOUNT HOLE MOUNT HOLE MOUNT MOUNT
L22 47 1.17 67144080 67144460 RL-5471-6 — PE-53822 PE-53822-S DO3316-473
L23 33 1.4 67144090 67144470 RL-5471-7 — PE-53823 PE-53823-S DO3316-333
L24 22 1.7 67148370 67144480 RL-1283-22-43 — PE-53824 PE-53824-S DO3316-223
L26 330 0.8 67144100 67144480 RL-5471-1 — PE-53826 PE-53826-S DO5022P-334
L27 220 1 67144110 67144490 RL-5471-2 — PE-53827 PE-53827-S DO5022P-224
L28 150 1.2 67144120 67144500 RL-5471-3 — PE-53828 PE-53828-S DO5022P-154
L29 100 1.47 67144130 67144510 RL-5471-4 — PE-53829 PE-53829-S DO5022P-104
L30 68 1.78 67144140 67144520 RL-5471-5 — PE-53830 PE-53830-S DO5022P-683
L35 47 2.15 67144170 — RL-5473-1 — PE-53935 PE-53935-S —
9.1.8 Output Voltage Ripple and Transients
The output voltage of a switching power supply operating in the continuous mode contains a sawtooth ripple
voltage at the switcher frequency, and may also contain short voltage spikes at the peaks of the sawtooth
waveform.
The output ripple voltage is a function of the inductor sawtooth ripple current and the ESR of the output
capacitor. A typical output ripple voltage can range from approximately 0.5% to 3% of the output voltage. To
obtain low ripple voltage, the ESR of the output capacitor must be low; however, caution must be exercised when
using extremely low ESR capacitors because they can affect the loop stability, resulting in oscillation problems. If
very low output ripple voltage is required (less than 20 mV), TI recommends a post ripple filter (see Figure 45).
The inductance required is typically between 1 μH and 5 μH, with low DC resistance, to maintain good load
regulation. A low ESR output filter capacitor is also required to assure good dynamic load response and ripple
reduction. The ESR of this capacitor may be as low as desired, because it is out of the regulator feedback loop.
Figure 27 shows a typical output ripple voltage, with and without a post ripple filter.
When observing output ripple with a scope, it is essential that a short, low inductance scope probe ground
connection be used. Most scope probe manufacturers provide a special probe terminator which is soldered onto
the regulator board, preferably at the output capacitor. This provides a very short scope ground, thus eliminating
the problems associated with the 3 inch ground lead normally provided with the probe, and provides a much
cleaner and more accurate picture of the ripple voltage waveform.
The voltage spikes are caused by the fast switching action of the output switch, the diode, the parasitic
inductance of the output filter capacitor, and its associated wiring. To minimize these voltage spikes, the output
capacitor must be designed for switching regulator applications, and the lead lengths must be kept very short.
Wiring inductance, stray capacitance, as well as the scope probe used to evaluate these transients, all contribute
to the amplitude of these spikes.
Figure 39. Peak-to-Peak Inductor
Ripple Current vs Load Current
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When a switching regulator is operating in the continuous mode, the inductor current waveform ranges from a
triangular to a sawtooth type of waveform (depending on the input voltage). For a given input and output voltage,
the peak-to-peak amplitude of this inductor current waveform remains constant. As the load current increases or
decreases, the entire sawtooth current waveform also rises and falls. The average value (or the center) of this
current waveform is equal to the DC load current.
If the load current drops to a low enough level, the bottom of the sawtooth current waveform reaches zero, and
the switcher smoothly changes from a continuous to a discontinuous mode of operation. Most switcher designs
(regardless how large the inductor value is) is forced to run discontinuous if the output is lightly loaded. This is a
perfectly acceptable mode of operation.
In a switching regulator design, knowing the value of the peak-to-peak inductor ripple current (ΔIIND) can be
useful for determining a number of other circuit parameters. Parameters such as, peak inductor or peak switch
current, minimum load current before the circuit becomes discontinuous, output ripple voltage and output
capacitor ESR can all be calculated from the peak-to-peak ΔIIND. When the inductor nomographs shown in
Figure 35 through Figure 38 are used to select an inductor value, the peak-to-peak inductor ripple current can
immediately be determined. Figure 39 shows the range of (ΔIIND) that can be expected for different load currents.
Figure 39 also shows how the peak-to-peak inductor ripple current (ΔIIND) changes as the designer goes from the
lower border to the upper border (for a given load current) within an inductance region. The upper border
represents a higher input voltage, while the lower border represents a lower input voltage (see Inductor Selection
Guides).
These curves are only correct for continuous mode operation, and only if the inductor selection guides are used
to select the inductor value
Consider the following example:
VOUT = 5 V, maximum load current of 800 mA
VIN = 12 V, nominal, varying between 10 V and 14 V.
The selection guide in Figure 36 shows that the vertical line for a 0.8-A load current and the horizontal line for the
12-V input voltage intersect approximately midway between the upper and lower borders of the 68-μH inductance
region. A 68-μH inductor allows a peak-to-peak inductor current (ΔIIND) to a percentage of the maximum load
current. Referring to Figure 39, follow the 0.8-A line approximately midway into the inductance region, and read
the peak-to-peak inductor ripple current (ΔIIND) on the left hand axis (approximately 300-mA p-p).
As the input voltage increases to 14 V, it approaches the upper border of the inductance region, and the inductor
ripple current increases. Figure 39 shows that for a load current of 0.8 A, the peak-to-peak inductor ripple current
(ΔIIND) is 300 mA with 12-V in, and can range from 340 mA at the upper border (14-V in) to 225 mA at the lower
border (10-V in).
Once the ΔIIND value is known, the following formulas can be used to calculate additional information about the
switching regulator circuit.
1. Peak Inductor or peak switch current
2. Minimum load current before the circuit becomes discontinuous
3. Output Ripple Voltage = (ΔIIND) × (ESR of COUT) = 0.3 A × 0.16 Ω = 48 mVp-p
4. ESR of COUT
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9.1.9 Open Core Inductors
Another possible source of increased output ripple voltage or unstable operation is from an open core inductor.
Ferrite bobbin or stick inductors have magnetic lines of flux flowing through the air from one end of the bobbin to
the other end. These magnetic lines of flux induce a voltage into any wire or PCB copper trace that comes within
the magnetic field of the inductor. The strength of the magnetic field, the orientation and location of the PC
copper trace to the magnetic field, and the distance between the copper trace and the inductor determine the
amount of voltage generated in the copper trace. Another way of looking at this inductive coupling is to consider
the PCB copper trace as one turn of a transformer (secondary) with the inductor winding as the primary. Many
millivolts can be generated in a copper trace located near an open core inductor, which can cause stability
problems or high output ripple voltage problems.
If unstable operation is seen, and an open core inductor is used, it is possible that the location of the inductor
with respect to other PC traces may be the problem. To determine if this is the problem, temporarily raise the
inductor away from the board by several inches and then check circuit operation. If the circuit now operates
correctly, then the magnetic flux from the open core inductor is causing the problem. Substituting a closed-core
inductor such as a torroid or E-core correct the problem, or re-arranging the PC layout may be necessary.
Magnetic flux cutting the IC device ground trace, feedback trace, or the positive or negative traces of the output
capacitor must be minimized.
Sometimes, placing a trace directly beneath a bobbin inductor provides good results, provided it is exactly in the
center of the inductor (because the induced voltages cancel themselves out). However, problems could arise if
the trace is off center. If flux problems are present, even the direction of the inductor winding can make a
difference in some circuits.
This discussion on open core inductors is not to frighten users, but to alert them on what kind of problems to
watch out for when using them. Open core bobbin or stick inductors are an inexpensive, simple way of making a
compact, efficient inductor, and they are used by the millions in many different applications.
Circuit Data for Temperature Rise Curve TO-220 Package (T)
Capacitors Through hole electrolytic
Inductor Through hole, Schott, 68 μH
Diode Through hole, 3-A, 40-V, Schottky
Printed-circuit board 3 square inches single sided 2 oz. copper (0.0028″)
Figure 40. Junction Temperature Rise, TO-220
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Circuit Data for Temperature Rise Curve DDPAK Package (S)
Capacitors Surface mount tantalum, molded D size
Inductor Surface mount, Schott, 68 μH
Diode Surface mount, 3-A, 40-V, Schottky
Printed-circuit board 3 square inches single sided 2 oz. copper (0.0028″)
Figure 41. Junction Temperature Rise, DDPAK
9.2 Typical Application
9.2.1 LM2598 Fixed Output Series Buck Regulator
Component Values shown are for VIN = 15 V, VOUT = 5 V, ILOAD = 1 A.
120-μF, 50-V, Aluminum Electrolytic Nichicon PL Series
120-μF, 35-V Aluminum Electrolytic, Nichicon PL Series
3-A, 40-V Schottky Rectifier, 1N5822
68-μH, L30
Typical Values
*CSS: — 0.1 μF
CDELAY: — 0.1 μF
RPull Up: — 4.7k
Figure 42. Fixed Output Voltage Version
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Typical Application (continued)
9.2.1.1 Design Requirements
Table 3 lists the design parameters of this application example.
Table 3. Design Parameters
PARAMETERS EXAMPLE VALUE
Regulated output voltage (3.3 V, 5 V or 12 V), VOUT 5 V
Maximum DC input voltage, VIN(max) 12 V
Maximum load current, ILOAD(max) 1 A
9.2.1.2 Detailed Design Procedure
9.2.1.2.1 Inductor Selection (L1)
1. Select the correct inductor value selection guide from Figure 35, Figure 36, or Figure 37 (Output voltages of
3.3 V, 5 V, or 12 V respectively.) Use the inductor selection guide for the 5-V version shown in Figure 36.
2. From the inductor value selection guide, identify the inductance region intersected by the maximum input
voltage line and the maximum load current line. Each region is identified by an inductance value and an
inductor code (LXX). From the inductor value selection guide shown in Figure 36, the inductance region
intersected by the 12-V horizontal line and the 1-A vertical line is 68 μH, and the inductor code is L30.
3. Select an appropriate inductor from the four manufacturer's part numbers listed in Table 2. The inductance
value required is 68 μH. See row L30 of Table 2 and choose an inductor part number from any of the four
manufacturers shown. (In most instance, both through hole and surface mount inductors are available.)
9.2.1.2.2 Output Capacitor Selection (COUT)
1. In the majority of applications, low ESR (Equivalent Series Resistance) electrolytic capacitors between 47 μF
and 330 μF and low ESR solid tantalum capacitors between 56 μF and 270 μF provide the best results. This
capacitor must be located close to the IC using short capacitor leads and short copper traces. Do not use
capacitors larger than 330 μF.
For additional information, see section on output capacitors in Output Capacitor (COUT) section.
2. To simplify the capacitor selection procedure, see Figure 38 for quick design component selection. This table
contains different input voltages, output voltages, and load currents, and lists various inductors and output
capacitors that provide the best design solutions.
From Figure 38, locate the 5-V output voltage section. In the load current column, choose the load current
line that is closest to the current required for the application; for this example, use the 1-A line. In the
maximum input voltage column, select the line that covers the input voltage required for the application; in
this example, use the 15-V line. The rest of this line shows the recommended inductors and capacitors that
provide the best overall performance.
The capacitor list contains both through hole electrolytic and surface mount tantalum capacitors from four
different capacitor manufacturers. TI recommends using both the manufacturers and the manufacturer's
series that are listed in Figure 38.
In this example aluminum electrolytic capacitors from several different manufacturers are available with the
range of ESR numbers required.
– 220-μF, 25-V Panasonic HFQ Series
– 220 μF, 25-V Nichicon PL Series
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Table 4. LM2598 Fixed Voltage Quick Design Component Selection Table
OUTPUT CAPACITOR
CONDITIONS INDUCTOR THROUGH-HOLE
ELECTROLYTIC SURFACE-MOUNT TANTALUM
OUTPUT LOAD MAX INPUT
VOLTAGE CURRENT VOLTAGE INDUCTANCE INDUCTOR
PANASONIC NICHICON AVX TPS SPRAGUE
(μH) (#) HFQ SERIES PL SERIES SERIES 595D SERIES(V) (A) (V) (μF/V) (μF/V) (μF/V) (μF/V)
5 22 L24 330/16 330/16 220/10 330/10
7 33 L23 270/25 270/25 220/10 270/10
1
10 47 L31 220/25 220/35 220/10 220/10
3.3 40 68 L30 180/35 220/35 220/10 180/10
6 47 L13 220/25 220/16 220/16 220/10
0.5 10 68 L21 150/35 150/25 100/16 150/16
40 100 L20 150/35 82/35 100/16 100/20
8 33 L28 330/16 330/16 220/10 270/10
10 47 L31 220/25 220/25 220/10 220/10
1
15 68 L30 180/35 180/35 220/10 150/16
5 40 100 L29 180/35 120/35 100/16 120/16
9 68 L21 180/16 180/16 220/10 150/16
0.5 20 150 L19 120/25 120/25 100/16 100/20
40 150 L19 100/25 100/25 68/20 68/25
15 47 L31 220/25 220/25 68/20 120/20
18 68 L30 180/35 120/25 68/20 120/20
1
30 150 L36 82/25 82/25 68/20 100/20
12 40 220 L35 82/25 82/25 68/20 68/25
15 68 L21 180/25 180/25 68/20 120/20
0.5 20 150 L19 82/25 82/25 68/20 100/20
40 330 L26 56/25 56/25 68/20 68/25
3. The capacitor voltage rating for electrolytic capacitors must be at least 1.5 times greater than the output
voltage, and often much higher voltage ratings are required to satisfy the low ESR requirements for low
output ripple voltage
For a 5-V output, a capacitor voltage rating at least 7.5 V or more is required. But, in this example, even a
low ESR, switching grade, 220-μF, 10-V aluminum electrolytic capacitor would exhibit approximately 225 mΩ
of ESR (see the curve in Figure 32 for the ESR vs voltage rating). This amount of ESR would result in
relatively high output ripple voltage. To reduce the ripple to 1% of the output voltage, or less, a capacitor with
a higher voltage rating (lower ESR) must be selected. A 16-V or 25-V capacitor reduces the ripple voltage by
approximately half.
9.2.1.2.3 Catch Diode Selection (D1)
1. The catch diode current rating must be at least 1.3 times greater than the maximum load current. Also, if the
power supply design must withstand a continuous output short, the diode must have a current rating equal to
the maximum current limit of the LM2598. The most stressful condition for this diode is an overload or
shorted output condition. See Table 5. In this example, a 3-A, 20-V, 1N5820 Schottky diode provides the
best performance, and does not overstressed even for a shorted output.
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Table 5. Diode Selection Table
1-A DIODES 3-A DIODES
VR SURFACE MOUNT THROUGH HOLE SURFACE MOUNT THROUGH HOLE
SCHOTTKY ULTRA FAST SCHOTTKY ULTRA FAST SCHOTTKY ULTRA FAST ULTRA FASTRECOVERY RECOVERY RECOVERY SCHOTTKY RECOVERY
SK12 1N5817 IN5820
20 V All of these SR102 All of these SK32 All of these SR302 All of these
diodes are rated diodes are rated diodes are rated diodes are rated
to at least 50 V. to at least 50 V. to at least 50 V. MBR320 to at least 50 V.
SK13 1N5818 1N5821
30 V MBRS130 SR103 SK33 MBR330
11DQ03 31DQ03
SK14 1N5822
MBRS140 1N5819 SK34 SR304
40 V
10BQ040 SR104 MBRS340 MBR340
10MQ040 MURS120 11DQ04 MUR120 30WQ04 MURS320 31DQ04 MUR320
50 V MBRS160 10BF10 SR105 SK35 30WF10 SR305 30WF10
or 10BQ050 MBR150 MBRS360 MBR350
more 10MQ060 11DQ05 30WQ05 31DQ05
2. The reverse voltage rating of the diode must be at least 1.25 times the maximum input voltage.
3. This diode must be fast (short reverse recovery time) and must be located close to the LM2598 using short
leads and short printed circuit traces. Because of their fast switching speed and low forward voltage drop,
Schottky diodes provide the best performance and efficiency, and must be the first choice, especially in low
output voltage applications. Ultra-fast recovery, or high-efficiency rectifiers also provide good results. Ultra-
fast recovery diodes typically have reverse recovery times of 50 ns or less. Rectifiers such as the 1N5400
must not be used because they are too slow.
9.2.1.2.4 Input Capacitor (CIN)
A low ESR aluminum or tantalum bypass capacitor is required between the input pin and ground to prevent large
voltage transients from appearing at the input. In addition, the RMS current rating of the input capacitor must be
selected to be at least ½ the DC load current. The capacitor manufacturers data sheet must be checked to
assure that this current rating is not exceeded. Figure 31 shows typical RMS current ratings for several different
aluminum electrolytic capacitor values.
This capacitor must be located close to the IC using short leads and the voltage rating must be approximately 1.5
times the maximum input voltage.
If solid tantalum input capacitors are used, TI recommends they be surge current tested by the manufacturer.
Use caution when using ceramic capacitors for input bypassing, because it may cause severe ringing at the VIN
pin.
The important parameters for the Input capacitor are the input voltage rating and the RMS current rating. With a
nominal input voltage of 12 V, an aluminum electrolytic capacitor with a voltage rating greater than 18 V
(1.5 × VIN) is necessary. The next higher capacitor voltage rating is 25 V.
The RMS current rating requirement for the input capacitor in a buck regulator is approximately ½ the DC load
current. In this example, with a 1-A load, a capacitor with a RMS current rating of at least 500 mA is required.
Figure 31 shows curves that can be used to select an appropriate input capacitor. From the curves, locate the
25-V line and note which capacitor values have RMS current ratings greater than 500 mA. Either a 180-μF or
220-μF, 25-V capacitor could be used.
For a through-hole design, a 220-μF, 25-V electrolytic capacitor (Panasonic HFQ series or Nichicon PL series or
equivalent) would be adequate. Other types or other manufacturers' capacitors can be used provided the RMS
ripple current ratings are adequate.
For surface-mount designs, solid tantalum capacitors are recommended. The TPS series available from AVX,
and the 593D series from Sprague are both surge current tested.
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9.2.1.3 Application Curves
Continuous Mode Switching Waveforms Load Transient Response for Continuous Mode
V = 20 V, V = 5 V, I = 1 A, L = 68 μH, VIN = 20 V, VOUT = 5 V, ILOAD = 250 mA to 750 mA,IN OUT LOAD
C = 120 μF, C ESR = 100 mΩ L = 68 μH, COUT = 120 μF, COUT ESR = 100 mΩOUT OUT
A: Output Pin Voltage, 10 V/div. A: Output Voltage, 100 mV/div. (AC)
B: Inductor Current 0.5 A/div. B: 250-mA to 750-mA Load Pulse
C: Output Ripple Voltage, 50 mV/div.
Figure 43. Horizontal Time Base: 2 μs/div Figure 44. Horizontal Time Base: 100 μs/div
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9.2.2 LM2598 Adjustable Output Series Buck Regulator
where VREF = 1.23 V
Select R1 to be approximately 1 kΩ, use a 1% resistor for best stability.
Component Values shown are for VIN = 20 V,
VOUT = 10 V, ILOAD = 1 A.
CIN — 120 μF, 35-V, Aluminum Electrolytic Nichicon PL Series
COUT — 120 μF, 35-V Aluminum Electrolytic, Nichicon PL Series
D1 —3-A, 40-V Schottky Rectifier, 1N5822
L1 —100 μH, L29
R1 —1 kΩ, 1%
R2 —7.1 kΩ, 1%
CFF — 3.3 nF, See Feedforward Capacitor (CFF)
RFF — 3 kΩ, See Feedforward Capacitor (CFF)
Typical Values
CSS—0.1 μF
CDELAY—0.1 μF
RPULL UP—4.7 kΩ
Figure 45. Adjustable Output Voltage Version
9.2.2.1 Design Requirements
Table 6 lists the design parameters for this application example.
Table 6. Design Parameters
PARAMETERS EXAMPLE VALUE
Regulated output voltage (3.3 V, 5 V or 12 V), VOUT 20 V
Maximum DC input voltage, VIN(max) 28 V
Maximum load current, ILOAD(max) 1 A
Switching frequency, F Fixed at a nominal 150 kHz
9.2.2.2 Detailed Design Procedure
9.2.2.2.1 Programming Output Voltage
Select R1 and R2, as shown in Figure 45.
Use Equation 1 to select the appropriate resistor values.
(1)
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Select a value for R1 with Equation 2 between 240 Ω and 1.5 kΩ. The lower resistor values minimize noise
pickup in the sensitive feedback pin. (For the lowest temperature coefficient and the best stability with time, use
1% metal film resistors.)
(2)
Select R1 with Equation 3 to be 1 kΩ, 1%. Solve for R2.
(3)
R2 = 1k (16.26 – 1) = 15.26k, closest 1% value is 15.4 kΩ.
R2 = 15.4 kΩ.
9.2.2.2.2 Inductor Selection (L1)
1. Calculate the inductor Volt • microsecond constant E • T (V • μs) with Equation 4.
where
• VSAT = internal switch saturation voltage = 1 V
• VD = diode forward voltage drop = 0.5 V (4)
Calculate the inductor Volt • microsecond constant (E • T) with Equation 5.
(5)
2. Use the E • T value from the previous formula and match it with the E • T number on the vertical axis of the
see the inductor selection graphs in Figure 35 through Figure 38.
E • T = 34.8 (V • μs)
3. On the horizontal axis, select the maximum load current.
ILOAD(max) = 1 A
4. Identify the inductance region intersected by the E • T value and the Maximum Load Current value. Each
region is identified by an inductance value and an inductor code (LXX).
From the inductor selection graphs in Figure 35 through Figure 38, the inductance region intersected by the
35 (V • μs) horizontal line and the 1-A vertical line is 100 μH, and the inductor code is L29.
5. Select an appropriate inductor from the four manufacturer's part numbers listed in Table 2.
From the table in Table 2, locate line L29, and select an inductor part number from the list of manufacturers'
part numbers.
9.2.2.2.3 Output Capacitor Selection (COUT)
1. In the majority of applications, low ESR electrolytic or solid tantalum capacitors between 82 μF and 220 μF
provide the best results. This capacitor must be located close to the IC using short capacitor leads and short
copper traces. Do not use capacitors larger than 220 μF. For additional information, see Output Capacitor
(COUT).
2. To simplify the capacitor selection procedure, see Table 1 for a quick design guide. This table contains
different output voltages, and lists various output capacitors that provide the best design solutions.
From Table 1, locate the output voltage column. From that column, locate the output voltage closest to the
output voltage in your application. In this example, select the 24-V line. Under the Output Capacitor (COUT)
section, select a capacitor from the list of through hole electrolytic or surface mount tantalum types from four
different capacitor manufacturers. TI recommends that both the manufacturers and the manufacturers series
that are listed in Table 1 be used.
In this example, through hole aluminum electrolytic capacitors from several different manufacturers are
available:
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– 82-μF, 35-V Panasonic HFQ Series
– 82-μF, 35-V Nichicon PL Series
3. The capacitor voltage rating must be at least 1.5 times greater than the output voltage, and often much
higher voltage ratings are required to satisfy the low ESR requirements required for low output ripple voltage.
For a 20-V output, a capacitor rating of at least 30 V or more is required. In this example, either a 35-V or
50-V capacitor would work. A 35-V rating was chosen although a 50-V rating could also be used if a lower
output ripple voltage is required.
Other manufacturers or other types of capacitors may also be used, provided the capacitor specifications
(especially the 100 kHz ESR) closely match the types listed in Table 1. Refer to the capacitor manufacturers
data sheet for this information.
9.2.2.2.4 Feedforward Capacitor (CFF)
For output voltages greater than approximately 10 V, an additional capacitor is required (use Equation 6; see
Figure 45). The compensation capacitor is typically between 50 pF and 10 nF, and is wired in parallel with the
output voltage setting resistor, R2. It provides additional stability for high output voltages, low input or output
voltages, or very low ESR output capacitors, such as solid tantalum capacitors.
(6)
This capacitor type can be ceramic, plastic, silver mica, etc. (Because of the unstable characteristics of ceramic
capacitors made with Z5U material, they are not recommended.)
The table shown in Table 1 contains feedforward capacitor values for various output voltages. In this example, a
1-nF capacitor is required.
9.2.2.2.5 Catch Diode Selection (D1)
1. The catch diode current rating must be at least 1.3 times greater than the maximum load current. Also, if the
power supply design must withstand a continuous output short, the diode must have a current rating equal to
the maximum current limit of the LM2598. The most stressful condition for this diode is an overload or
shorted output condition.
See Table 5. Schottky diodes provide the best performance, and in this example a 3-A, 40-V, 1N5822
Schottky diode is a good choice. The 3-A diode rating is more than adequate and does not overstressed
even for a shorted output.
2. The reverse voltage rating of the diode must be at least 1.25 times the maximum input voltage.
3. This diode must be fast (short reverse recovery time) and must be placed close to the LM2598 using short
leads and short printed circuit traces. Because of their fast switching speed and low forward voltage drop,
Schottky diodes provide the best performance and efficiency, and must be the first choice, especially in low
output voltage applications. Ultra-fast recovery or high-efficiency rectifiers are also good choices, but some
types with an abrupt turnoff characteristic may cause instability or EMl problems. Ultra-fast recovery diodes
typically have reverse recovery times of 50 ns or less. Rectifiers such as the 1N4001 series must not be
used because they are too slow.
9.2.2.2.6 Input Capacitor (CIN)
A low ESR aluminum or tantalum bypass capacitor is required between the input pin and ground to prevent large
voltage transients from appearing at the input. In addition, the RMS current rating of the input capacitor must be
selected to be at least ½ the DC load current. The capacitor manufacturers data sheet must be checked to
assure that this current rating is not exceeded. Figure 31 shows typical RMS current ratings for several different
aluminum electrolytic capacitor values.
This capacitor must be located close to the IC using short leads and the voltage rating must be approximately 1.5
times the maximum input voltage.
If solid tantalum input capacitors are used, it is recomended that they be surge current tested by the
manufacturer.
Use caution when using a high dielectric constant ceramic capacitor for input bypassing, because it may cause
severe ringing at the VIN pin.
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The important parameters for the input capacitor are the input voltage rating and the RMS current rating. With a
nominal input voltage of 28 V, an aluminum electrolytic aluminum electrolytic capacitor with a voltage rating
greater than 42 V (1.5 × VIN) is required. Because the the next higher capacitor voltage rating is 50 V, a 50-V
capacitor must be used. The capacitor voltage rating of (1.5 × VIN) is a conservative guideline, and can be
modified somewhat if desired.
The RMS current rating requirement for the input capacitor of a buck regulator is approximately ½ the DC load
current. In this example, with a 1-A load, a capacitor with a RMS current rating of at least 500 mA is required.
Figure 31 shows curves that can be used to select an appropriate input capacitor. From the curves, locate the
50-V line and note which capacitor values have RMS current ratings greater than 500 mA. Either a 100-μF or
120-μF, 50-V capacitor could be used.
For a through-hole design, a 120-μF, 50-V electrolytic capacitor (Panasonic HFQ series or Nichicon PL series or
equivalent) would be adequate. Other types or other manufacturers' capacitors can be used provided the RMS
ripple current ratings are adequate.
For surface-mount designs, solid tantalum capacitors can be used, but caution must be exercised with regard to
the capacitor surge current rating (see Input Capacitor (CIN)). The TPS series available from AVX, and the 593D
series from Sprague are both surge current tested.
9.2.2.3 Application Curves
Load Transient Response for Discontinuous Mode
VIN = 20 V, VOUT = 5 V, ILOAD = 250 mA to 750 mA,
L = 22 μH, COUT = 220 μF, COUT ESR = 50 mΩ
A: Output Voltage, 100 mV/div. (AC)
Discontinuous Mode Switching Waveforms
VIN = 20 V,VOUT = 5 V, I
B: 250-mA to 750-mA Load Pulse
LOAD = 600 mA, L = 22 μH,
COUT = 220 μF,COUT ESR = 50 mΩ
A: Output Pin Voltage, 10 V/div.
B: Inductor Current 0.5 A/div.
C: Output Ripple Voltage, 50 mV/div.
Figure 46. Horizontal Time Base: 2 μs/div Figure 47. Horizontal Time Base: 200 μs/div
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10 Power Supply Recommendations
The LM2598 is designed to operate from an input voltage supply up to 40 V. This input supply must be well
regulated and able to withstand maximum input current and maintain a stable voltage.
11 Layout
11.1 Layout Guidelines
As in any switching regulator, layout is very important. Rapidly switching currents associated with wiring
inductance can generate voltage transients which can cause problems. For minimal inductance and ground
loops, the wires indicated by heavy lines must be wide printed circuit traces and must be kept as short as
possible. For best results, external components must be placed as close to the switcher lC as possible using
ground plane construction or single point grounding.
If open core inductors are used, take special care regarding the location and positioning of this type of inductor.
Allowing the inductor flux to intersect sensitive feedback, lC groundpath and COUT wiring can cause problems.
When using the adjustable version, special care must be taken as to the location of the feedback resistors and
the associated wiring. Physically place both resistors near the IC, and route the wiring away from the inductor,
especially an open core type of inductor (see Open Core Inductors for more information).
11.2 Layout Examples
CIN—150-μF, 50-V Aluminum Electrolytic, Panasonic HFQ series
COUT—120-μF, 25-V Aluminum Electrolytic, Panasonic HFQ series
D1 — 3-A, 40-V Schottky Rectifier, 1N5822
L1 — 68-μH, L30, Renco, Through hole
RPULL-UP — 10 kΩ
CDELAY — 0.1 μF
CSD/SS — 0.1 μF
Figure 48. Typical Through-Hole PCB Layout, Fixed Output (1x Size), Double-Sided, Through-Hole Plated
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Layout Examples (continued)
CIN — 150-μF, 50-V, Aluminum Electrolytic, Panasonic HFQ series
COUT — 120-μF, 25-V Aluminum Electrolytic, Panasonic HFQ series
D1 — 3-A, 40-V Schottky Rectifier, 1N5822
L1 — 68-μH, L30, Renco, Through hole
R1 — 1 kΩ, 1%
R2—Use formula in Design Procedure
CFF—See Feedforward Capacitor (CFF).
RFF—See Feedforward Capacitor (CFF).
RPULL-UP—10 kΩ
CDELAY — 0.1-μF
CSD/SS — 0.1 μF
Figure 49. Typical Through-Hole PCB Layout, Adjustable Output (1x Size), Double-Sided, Through-Hole
Plated
11.3 Thermal Considerations
The LM2598 is available in two packages: a 7-pin TO-220 (T) and a 7-pin surface mount DDPAK (S).
The TO-220 package can be used without a heat sink for ambient temperatures up to approximately 50°C
(depending on the output voltage and load current). Figure 40 shows the LM2598T junction temperature rises
above ambient temperature for different input and output voltages. The data for these curves was taken with the
LM2598T (TO-220 package) operating as a switching regulator in an ambient temperature of 25°C (still air).
These temperature rise numbers are all approximate and there are many factors that can affect these
temperatures. Higher ambient temperatures require some heat sinking, either to the PCB or a small external heat
sink.
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Thermal Considerations (continued)
The DDPAK surface mount package tab is designed to be soldered to the copper on a printed-circuit board
(PCB). The copper and the board are the heat sink for this package and the other heat producing components,
such as the catch diode and inductor. The PCB copper area that the package is soldered to must be at least 0.4
in2, and ideally must have 2 or more square inches of 2 oz. (0.0028) in) copper. Additional copper area improves
the thermal characteristics, but with copper areas greater than approximately 3 in2, only small improvements in
heat dissipation are realized. If further thermal improvements are required, TI recommends double-sided or
multilayer PCB with large copper areas.
Figure 41 shows the LM2598S (DDPAK package) junction temperature rise above ambient temperature with a 1-
A load for various input and output voltages. This data was taken with the circuit operating as a buck switching
regulator with all components mounted on a PCB to simulate the junction temperature under actual operating
conditions. This curve can be used for a quick check for the approximate junction temperature for various
conditions, but be aware that there are many factors that can affect the junction temperature.
For the best thermal performance, wide copper traces and generous amounts of PCB copper must be used in
the board layout. (One exception to this is the output (switch) pin, which must not have large areas of copper.)
Large areas of copper provide the best transfer of heat (lower thermal resistance) to the surrounding air, and
moving air lowers the thermal resistance even further.
Package thermal resistance and junction temperature rise numbers are all approximate, and there are many
factors that affect these numbers. Some of these factors include board size, shape, thickness, position, location,
and even board temperature. Other factors are trace width, total printed-circuit copper area, copper thickness,
single- or double-sided multilayer board, and the amount of solder on the board. The effectiveness of the PCB to
dissipate heat also depends on the size, quantity, and spacing of other components on the board, as well as
whether the surrounding air is still or moving. Furthermore, some of these components such as the catch diode
adds heat to the PCB and the heat can vary as the input voltage changes. For the inductor, depending on the
physical size, type of core material, and the DC resistance, it could either act as a heat sink taking heat away
from the board, or it could add heat to the board.
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12 Device and Documentation Support
12.1 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.2 Trademarks
E2E is a trademark of Texas Instruments.
SIMPLE SWITCHER is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.3 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.4 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
www.ti.com 15-Feb-2016
PACKAGING INFORMATION
Orderable Device Status Package Type Package Pins Package Eco Plan Lead/Ball Finish MSL Peak Temp Op Temp (°C) Device Marking Samples
(1) Drawing Qty (2) (6) (3) (4/5)
LM2598S-12/NOPB ACTIVE DDPAK/ KTW 7 45 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -12 P+
LM2598S-3.3/NOPB ACTIVE DDPAK/ KTW 7 45 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -3.3 P+
LM2598S-5.0 NRND DDPAK/ KTW 7 45 TBD Call TI Call TI -40 to 125 LM2598S
TO-263 -5.0 P+
LM2598S-5.0/NOPB ACTIVE DDPAK/ KTW 7 45 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -5.0 P+
LM2598S-ADJ/NOPB ACTIVE DDPAK/ KTW 7 45 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -ADJ P+
LM2598SX-12/NOPB ACTIVE DDPAK/ KTW 7 500 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -12 P+
LM2598SX-3.3/NOPB ACTIVE DDPAK/ KTW 7 500 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -3.3 P+
LM2598SX-5.0 NRND DDPAK/ KTW 7 500 TBD Call TI Call TI -40 to 125 LM2598S
TO-263 -5.0 P+
LM2598SX-5.0/NOPB ACTIVE DDPAK/ KTW 7 500 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -5.0 P+
LM2598SX-ADJ/NOPB ACTIVE DDPAK/ KTW 7 500 Pb-Free (RoHS CU SN Level-3-245C-168 HR -40 to 125 LM2598S
TO-263 Exempt) -ADJ P+
LM2598T-12/NOPB ACTIVE TO-220 NDZ 7 45 Green (RoHS CU SN Level-1-NA-UNLIM -40 to 125 LM2598T
& no Sb/Br) -12 P+
LM2598T-3.3/NOPB ACTIVE TO-220 NDZ 7 45 Green (RoHS CU SN Level-1-NA-UNLIM -40 to 125 LM2598T
& no Sb/Br) -3.3 P+
LM2598T-5.0/NOPB ACTIVE TO-220 NDZ 7 45 Green (RoHS CU SN Level-1-NA-UNLIM -40 to 125 LM2598T
& no Sb/Br) -5.0 P+
LM2598T-ADJ/NOPB ACTIVE TO-220 NDZ 7 45 Green (RoHS CU SN Level-1-NA-UNLIM -40 to 125 LM2598T
& no Sb/Br) -ADJ P+
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com 15-Feb-2016
(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 2
PACKAGE MATERIALS INFORMATION
www.ti.com 15-Feb-2016
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package Package Pins SPQ Reel Reel A0 B0 K0 P1 W Pin1
Type Drawing Diameter Width (mm) (mm) (mm) (mm) (mm) Quadrant
(mm) W1 (mm)
LM2598SX-12/NOPB DDPAK/ KTW 7 500 330.0 24.4 10.75 14.85 5.0 16.0 24.0 Q2
TO-263
LM2598SX-3.3/NOPB DDPAK/ KTW 7 500 330.0 24.4 10.75 14.85 5.0 16.0 24.0 Q2
TO-263
LM2598SX-5.0 DDPAK/ KTW 7 500 330.0 24.4 10.75 14.85 5.0 16.0 24.0 Q2
TO-263
LM2598SX-5.0/NOPB DDPAK/ KTW 7 500 330.0 24.4 10.75 14.85 5.0 16.0 24.0 Q2
TO-263
LM2598SX-ADJ/NOPB DDPAK/ KTW 7 500 330.0 24.4 10.75 14.85 5.0 16.0 24.0 Q2
TO-263
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com 15-Feb-2016
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM2598SX-12/NOPB DDPAK/TO-263 KTW 7 500 367.0 367.0 45.0
LM2598SX-3.3/NOPB DDPAK/TO-263 KTW 7 500 367.0 367.0 45.0
LM2598SX-5.0 DDPAK/TO-263 KTW 7 500 367.0 367.0 45.0
LM2598SX-5.0/NOPB DDPAK/TO-263 KTW 7 500 367.0 367.0 45.0
LM2598SX-ADJ/NOPB DDPAK/TO-263 KTW 7 500 367.0 367.0 45.0
Pack Materials-Page 2
MECHANICAL DATA
NDZ0007B
TA07B (Rev E)
www.ti.com
MECHANICAL DATA
KTW0007B
TS7B (Rev E)
BOTTOM SIDE OF PACKAGE
www.ti.com
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Copyright © 2016, Texas Instruments Incorporated
2N7000G
Small Signal MOSFET
200 mAmps, 60 Volts
N−Channel TO−92
Features http://onsemi.com
• AEC Qualified
• PPAP Capable 200 mAMPS
• This is a Pb−Free Device* 60 VOLTS
RDS(on) = 5
N−Channel
MAXIMUM RATINGS D
Rating Symbol Value Unit
Drain Source Voltage VDSS 60 Vdc
Drain−Gate Voltage (RGS = 1.0 M) VDGR 60 Vdc G
Gate−Source Voltage
− Continuous VGS ±20 Vdc S
− Non−repetitive (tp ≤ 50 s) VGSM ±40 Vpk
Drain Current mAdc
− Continuous ID 200
− Pulsed IDM 500 TO−92
Total Power Dissipation @ TC = 25°C PD 350 mW CASE 29
Derate above 25°C 2.8 mW/°C STYLE 22
Operating and Storage Temperature TJ, Tstg −55 to +150 °C
Range 1 12 2
3 3
THERMAL CHARACTERISTICS STRAIGHT LEAD BENT LEAD
Characteristic Symbol Max Unit BULK PACK TAPE & REEL
AMMO PACK
Thermal Resistance, Junction−to−Ambient RJA 357 °C/W
Maximum Lead Temperature for TL 300 °C MARKING DIAGRAM
Soldering Purposes, 1/16″ from case AND PIN ASSIGNMENT
for 10 seconds
Stresses exceeding Maximum Ratings may damage the device. Maximum
Ratings are stress ratings only. Functional operation above the Recommended 2N
Operating Conditions is not implied. Extended exposure to stresses above the 7000
Recommended Operating Conditions may affect device reliability. AYWW
1 3
Source Drain
2
Gate
A = Assembly Location
Y = Year
WW = Work Week
= Pb−Free Package
(Note: Microdot may be in either location)
ORDERING INFORMATION
See detailed ordering and shipping information in the package
dimensions section on page 2 of this data sheet.
*For additional information on our Pb−Free strategy and soldering details, please
download the ON Semiconductor Soldering and Mounting Techniques
Reference Manual, SOLDERRM/D.
© Semiconductor Components Industries, LLC, 2011 1 Publication Order Number:
April, 2011 − Rev. 8 2N7000/D
2N7000G
ELECTRICAL CHARACTERISTICS (TC = 25°C unless otherwise noted)
Characteristic Symbol Min Max Unit
OFF CHARACTERISTICS
Drain−Source Breakdown Voltage (VGS = 0, ID = 10 Adc) V(BR)DSS 60 − Vdc
Zero Gate Voltage Drain Current IDSS
(VDS = 48 Vdc, VGS = 0) − 1.0 Adc
(VDS = 48 Vdc, VGS = 0, TJ = 125°C) − 1.0 mAdc
Gate−Body Leakage Current, Forward (VGSF = 15 Vdc, VDS = 0) IGSSF − −10 nAdc
ON CHARACTERISTICS (Note 1)
Gate Threshold Voltage (VDS = VGS, ID = 1.0 mAdc) VGS(th) 0.8 3.0 Vdc
Static Drain−Source On−Resistance rDS(on)
(VGS = 10 Vdc, ID = 0.5 Adc) − 5.0
(VGS = 4.5 Vdc, ID = 75 mAdc) − 6.0
Drain−Source On−Voltage VDS(on) Vdc
(VGS = 10 Vdc, ID = 0.5 Adc) − 2.5
(VGS = 4.5 Vdc, ID = 75 mAdc) − 0.45
On−State Drain Current (VGS = 4.5 Vdc, VDS = 10 Vdc) Id(on) 75 − mAdc
Forward Transconductance (VDS = 10 Vdc, ID = 200 mAdc) gfs 100 − mhos
DYNAMIC CHARACTERISTICS
Input Capacitance Ciss − 60 pF
Output Capacitance (VDS = 25 V, VGS = 0, Coss − 25f = 1.0 MHz)
Reverse Transfer Capacitance Crss − 5.0
SWITCHING CHARACTERISTICS (Note 1)
Turn−On Delay Time (VDD = 15 V, ID = 500 mA, ton − 10 ns
Turn−Off Delay Time RG = 25 , RL = 30 , Vgen = 10 V) toff − 10
1. Pulse Test: Pulse Width ≤ 300 s, Duty Cycle ≤ 2.0%.
ORDERING INFORMATION
Device Package Shipping†
2N7000G TO−92 1000 Units / Bulk
(Pb−Free)
2N7000RLRAG TO−92 2000 Tape & Reel
(Pb−Free)
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specifications Brochure, BRD8011/D.
http://onsemi.com
2
2N7000G
2.0 1.0
1.8 TA = 25°C VDS = 10 V
-55°C 25°C
1.6 VGS = 10 V 0.8 125°C
1.4 9 V
1.2 8 V 0.6
1.0
7 V
0.8 0.4
0.6 6 V
0.4 5 V 0.2
0.2 4 V
3 V
0
0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10 0 1.0 2.0 3.0 4.0 5.0 6.0 7.0 8.0 9.0 10
VDS, DRAIN SOURCE VOLTAGE (VOLTS) VGS, GATE SOURCE VOLTAGE (VOLTS)
Figure 1. Ohmic Region Figure 2. Transfer Characteristics
2.4 1.2
2.2 1.05
2.0 VGS = 10 V
V
1.1 DS
= VGS
ID = 200 mA ID = 1.0 mA
1.8 1.10
1.6 1.0
1.4 0.95
1.2 0.9
1.0 0.85
0.8 0.8
0.6 0.75
0.4 0.7
-60 -20 +20 +60 +100 +140 -60 -20 +20 +60 +100 +140
T, TEMPERATURE (°C) T, TEMPERATURE (°C)
Figure 3. Temperature versus Static Figure 4. Temperature versus Gate
Drain−Source On−Resistance Threshold Voltage
http://onsemi.com
3
rDS(on), STATIC DRAIN-SOURCE ON-RESISTANCE
(NORMALIZED) ID, DRAIN CURRENT (AMPS)
VGS(th), THRESHOLD VOLTAGE (NORMALIZED) ID, DRAIN CURRENT (AMPS)
2N7000G
PACKAGE DIMENSIONS
TO−92 (TO−226)
CASE 29−11
ISSUE AM
A NOTES:B STRAIGHT LEAD 1. DIMENSIONING AND TOLERANCING PER ANSI
BULK PACK Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
R 3. CONTOUR OF PACKAGE BEYOND DIMENSION RIS UNCONTROLLED.
P 4. LEAD DIMENSION IS UNCONTROLLED IN P ANDBEYOND DIMENSION K MINIMUM.
L
SEATING INCHES MILLIMETERS
PLANE K DIM MIN MAX MIN MAX
A 0.175 0.205 4.45 5.20
B 0.170 0.210 4.32 5.33
C 0.125 0.165 3.18 4.19
D 0.016 0.021 0.407 0.533
X X D G 0.045 0.055 1.15 1.39
G H 0.095 0.105 2.42 2.66
J 0.015 0.020 0.39 0.50
H J K 0.500 --- 12.70 ---
L 0.250 --- 6.35 ---
V C N 0.080 0.105 2.04 2.66
P --- 0.100 --- 2.54
SECTION X−X R 0.115 --- 2.93 ---
1 N V 0.135 --- 3.43 ---
N
A NOTES:
R B BENT LEAD 1. DIMENSIONING AND TOLERANCING PER
TAPE & REEL ASME Y14.5M, 1994.
2. CONTROLLING DIMENSION: MILLIMETERS.
AMMO PACK 3. CONTOUR OF PACKAGE BEYOND
DIMENSION R IS UNCONTROLLED.
P 4. LEAD DIMENSION IS UNCONTROLLED IN PAND BEYOND DIMENSION K MINIMUM.
T
SEATING K MILLIMETERSPLANE DIM MIN MAX
A 4.45 5.20
B 4.32 5.33
C 3.18 4.19
D D 0.40 0.54X X G 2.40 2.80
G J 0.39 0.50
J K 12.70 ---
N 2.04 2.66
V
C P 1.50 4.00
R 2.93 ---
SECTION X−X V 3.43 ---
1 N STYLE 22:
PIN 1. SOURCE
2. GATE
3. DRAIN
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4